Natural fiber span reflectometer providing a spread spectrum virtual sensing array capability

ABSTRACT

A CW lightwave modulated by a continuously reiterated autocorrelated spectrum-spreading signal is launched into an end of a span of ordinary optical fiber cable. Portions of this lightwave back propagate to the launch end from a continuum of span locations because of innate fiber properties including Rayleigh effects. This is picked off the launch end and heterodyned producing an r.f. beat signal. The beat signal is processed by a plurality (can be thousands) of multifunction despreader, autocorrelator and de-multiplexer units respectively operated in different time delayed relationships to the timing base of launch signal reiteration. This provides r.f. time-domain reflectometry outputs representative of acoustic, or other signals incident upon virtual sensors at positions along the fiber corresponding to the various delay relationships. Material attenuation of undesired noises (e.g., reflections due to presence of couplers in the fiber cable line) is effected by the spectrum spreading and de-spreading.

Applicant claims the benefit of a provisional application, No.60/602,155 which was filed on 17 Aug. 2004, and which is entitled “FiberSpan-Based Rayleigh Effect Sensor System with Advance LocalizationCapability” by Robert M. Payton.

STATEMENT OF GOVERNMENT INTEREST

The invention described herein may be manufactured and used by or forthe Government of the United States of America for governmental purposeswithout the payment of any royalties thereon or therefore.

CROSS-REFERENCE TO RELATED APPLICATIONS

“Natural Fiber Span Reflectometer Providing a Virtual Signal SensingArray Capability” (Navy Case No 96517) filed on even date herewith inthe name of Robert M. Payton, hereby incorporated herein by reference inits entirety.

“Natural Fiber Span Reflectometer Providing A Virtual Phase SignalSensing Array Capability” (Navy Case No. 96518) filed on even dateherewith in the name of Robert M. Payton, hereby incorporated herein byreference in its entirety.

“Natural Fiber Span Reflectometer Providing a Virtual DifferentialSignal Sensing Array Capability” (Navy Case No. 96519) filed on evendate herewith in the name of Robert. M. Payton, hereby incorporatedherein by reference in its entirety.

BACKGROUND OF THE INVENTION

(1) Field of the Invention

The present invention relates generally to the field of time-domainreflectometers. More specifically, it relates to such reflectometerswhich are a part of a photonic system application in which the object ofthe reflectometry is a span of fiber which has an interrogation signallaunch end and a remote end. The invention enables the provision of alinear array of virtual sensors along the span. One particular type ofapplication toward which the invention is directed are acoustic securityalarm systems in which the span serves as a perimeter intrusionmonitoring line.

(2) Description of the Prior Art

The U.S. Department of the Navy has been engaged in the development oftowed acoustic arrays which are reflectometric systems in which theobject of the reflectometry is a fiber span having an interrogationsignal launch end and a remote end. One such development involvesforming a towed array of acoustic sensors along the span by the costlyprocess of irradiating Bragg reflective gratings into the fiber cable.These reflective gratings form the array of sensors of the reflectometryscheme of these systems. These towed arrays have a length of the orderof at least 1.0 km, and the need to irradiate the fiber has resulted inthe fiber spans costing hundreds of thousands of dollars each.

The Department of the Navy development activities have been furthertasked to apply their creative efforts to homeland defense problems. Aspart of this effort there is under consideration the use of areflectometer in which a fiber span is the object of the reflectometry.In this scheme, the fiber span provided with acoustic sensors would beused as an intrusion detector to monitor the perimeter of an areadesired to be secure. The span lengths for this type of applicationinclude lengths of the order of 5 km, (links of a U.S. border protectionnetwork, oil line protection, chemical plant protection, etc.). In suchperimeter monitoring applications thousands of acoustic sensors would berequired along the fiber span.

The cost of manufacturing such perimeter monitoring spans employingreflective Bragg grating sensors has been an obstacle to their use inperimeter intrusion monitoring applications. Thus, there is considerableinterest in the development of a reflectometer system in which a fiberspan is the object of the reflectometry optic array that does notrequire the high cost of Bragg reflective acoustic sensors.

Previous effort in solving related problems are described by thefollowing patents:

U.S. Pat. No. 5,194,847 issued Mar. 16, 1993 to H. Taylor and C. Leediscloses an apparatus for sensing intrusion into a predefined perimeterwhich comprises means for producing a coherent pulsed light, which isinjected into an optical sensing fiber having a first predeterminedlength and positioned along the predefined perimeter. A backscatteredlight in response to receiving the coherent light pulses is produced andcoupled into an optical receiving fiber. The backscattered light isdetected by a photodetector and a signal indicative of the backscatteredlight is produced. An intrusion is detectable from the produced signalas indicated by a change in the backscattered light. To increase thesensitivity of the apparatus, a reference fiber and interferometer mayalso be employed.

U.S. Pat. No. 6,285,806 issued on Sep. 4, 2001 to A. Kersey et al.,discloses an apparatus and method for measuring strain in an opticalfiber using the spectral shift of Rayleigh scattered light. Theinterference pattern produced by an air gap reflector and backscatterradiation is measured. Using Fourier Transforms, the spectrum of anysection of fiber can be extracted. Cross correlation with an unstrainedmeasurement produces a correlation peak. The location of the correlationpeak indicates the strain level in the selected portion of opticalfiber.

The above patents do not show how to obtain signals representingacoustic pressure signals incident upon a fiber span (to detectperimeter intrusion) at a very large number of sensing stations withoutinvolving high manufacturing costs. Consequently, those skilled in thearts will appreciate the present invention which addresses these andother problems.

SUMMARY OF THE INVENTION

The objects of the present invention include the provision of:

(1) A time-domain reflectometer wherein an optical fiber span is theobject of the reflectometry, and which provides output signalsrepresentative of acoustic pressure waves incident the span solely byvirtue of the natural, or innate, properties of commercial grade opticalfiber cables.

(2) The reflectometer described in object number (1), above, capable ofproviding acoustic wave signal sensing lengths of 5.0 km or more.

(3) The reflectometer described in object number (2), above, whichfacilitates the provision of a very large plurality (e.g. 5,000 or more)virtual acoustic sensors along the span.

(4) The reflectometer described in object number (1), above having amode of operation which inherently attenuates undesired noises due tospan line discontinuities, such as reflections caused by fiber cablecouplings.

(5) The reflectometer described in objects numbered (1) through (4),above, having special utility as a perimeter intrusion monitoring linefor an acoustic security alarm system.

(6) The reflectometer described in object numbered (1), above, which iscapable of providing output signals in the form of a phase signal whichvaries linearly with the acoustic pressure wave.

(7) The reflectometer described in object numbered (3), above, which iscapable of providing output signals in the form of phase differentialsignals across pairs of the virtual sensors.

(8) The reflectometer described in the object number (7), above,providing a capability of programmably selecting a pair, or pairs, ofvirtual acoustic sensors across which the phase signals are picked off,from among the plurality of virtual signals along the span.

These and other objects, features, and advantages of the presentinvention will become apparent from the drawings, the descriptions givenherein, and the appended claims. However, it will be understood that theabove listed objects and advantages of the invention are intended onlyas an aid in understanding aspects of the invention, and not intended tolimit the invention in any way, and do not form a comprehensive list ofobjects, features, and advantages.

Accordingly, a time-domain reflectometer is provided for sensing andproviding output signals representative of acoustic wave signalsincident on the fiber span which is the object of the reflectometry,wherein the innate properties of low cost, commercially available fiberoptic cables are employed to create a plurality (upwardly extending tovery large numbers, e.g., 5000 and more) virtual sensors.

The present invention is implemented as follows: Time and spatial domainmultiplexing and de-multiplexing of optical signals is accomplished byan electronic-delay or time of-transversal-delay coupled withmodulated-retransmission of a master or reference carrier wave. Eachindividual optical signal occupies a unique time-delay slot or bin. Amaster or carrier wave is modulated with each individual optical signaland delayed by the appropriate time interval specific to a particularsignal. All such signals are combined and simultaneously transmitted asa composite optical signal to a receiver where these are collected andphotodetected. By correlating the photodetected composite optical signalwith the master or reference carrier wave, each individual opticalsignal is sorted or demultiplexed into separate electronic signalchannels. The continuous wave nature of the master or reference carrierwave provides more power than a pulsed optical wave and heterodyneoptical reception of the invention allows a very low optical detectionthreshold or noise floor. The invention provides significant improvementover other systems because the optical noise floor is loweredconsiderably over more conventional means.

The invention applies to several applications. The invention allowsaudio bandwidth (tens of kilohertz bandwidth) providing time-domainreflectometry measurements of fiber optical cables or other opticalmediums such as glass, air, water, etc. Other time-domain reflectometrymethods do not sample the optical medium fast enough to detect tens ofkilohertz bandwidth variations in the medium. The invention also relatesto fiber optic sensors and optical sensors generally. A fiber opticsensor array is typically time-domain multiplexed by thetime-of-transversal of an interrogation lightwave to each sensor andback to a common optical collection and detection point. The inventionrelates generally to both amplitude and phase type optical senor arrays.The invention is an enabling technology for a Department of Navydevelopment known as the Rayleigh Optical Scattering and Encoding (ROSE)sensor system. The spatial separation of segmentation of a ROSE acousticarray into spatial channels is enabled by the invention.

The invention relates to acoustic security alarm systems, Naval towedarrays for sensing underwater acoustic signals, fiber optic buggingdevices, and many other potential ROSE applications. The invention alsorelates to non-fiber optical sensors such as: laser velocimeters; lasersimagers; laser radar; laser rangers; and remote laser acoustic, strain,motion or temperature measurement devices.

BRIEF DESCRIPTION OF THE DRAWINGS

A more complete understanding of the invention and many of the attendantadvantages thereto will be readily appreciated as the same becomesbetter understood by reference to the following detailed descriptionwhen considered in conjunction with the accompanying drawing, whereinlike reference numerals refer to like parts and wherein:

FIG. 1 is a graphical depiction of certain underlying physicalmechanisms of polarization;

FIG. 2 is a block diagram helpful in understanding the concept of launchof an interrogation signal along an optical fiber span providing avirtual array of pressure wave sensors by retrieval of backscatter fromRayleigh Optical Scatter (ROS) effects;

FIG. 3 is a block diagram of a natural fiber span time-domainreflectometer system in accordance with the present invention;

FIG. 4 is an electrical schematic of a balanced heterodyne opticdetector circuit;

FIG. 5 is an electrical schematic of an alternative embodiment of aphotodetector type heterodyner;

FIG. 6 is a block diagram of a programmable correlator subsystem, whichenables spatial sampling of optical signals on the fiber optic span ofthe system of FIG. 3, in order to provide a virtual array of acousticwave sensors therealong;

FIG. 7 is a block diagram depiction of a set of phase demodulatorcircuit assemblies which receives the outputs of the programmablecorrelator subsystem of FIG. 6;

FIG. 8 is a block diagram of one of the phase demodulator circuitassemblies of the set of FIG. 7;

FIG. 9 is a block diagram disclosing details of an I & Q demodulatorcomponent in a phase demodulator circuit assembly of FIG. 8;

FIG. 10 is a block diagram disclosing details of a digital embodiment ofthe phase detector component phase demodulator circuit assembly of FIG.8;

FIG. 11 is a block diagram disclosing details of an analog embodiment ofthe phase detector component phase demodulator circuit assembly of FIG.8;

FIG. 12 is a block diagram of a programmable routing and phase signalswitching network which provides selective pairing of the outputs of theset of phase demodulators of FIG. 7 to provide differential phasesignals across pairs of virtual sensors along the fiber span inaccordance with the present invention; and

FIG. 13 is a diagrammatic depiction of embodiment of invention of FIG. 3in which portions of the optical fiber span are wound around a hollowmandrel.

DESCRIPTION OF THE PREFERRED EMBODIMENT

(1) Description of Underlying Theories

a. Heterodyne Optical Detection

Optical receivers are built around photodetectors which detect opticalpower rather than instantaneous electric field. Typically thephotodetector output current is proportional to the incident opticalpower. This relationship severely limits the dynamic range of anincoherent optical receiver because for every decibel of optical powerlost in a receiver system two decibels of receiver output current islost. The square law characteristics of photodetectors limits typicalincoherent optical receivers (often called video detection receivers) todynamic ranges of less than 80 dB and optical detection noise floors togreater than −80 dBm per Hertz bandwidth. As illustration, suppose anelectric field E_(s)(t) [volt/meter] immersed in a material of impedanceη [Ohms] impinges upon a photodetector of responsivity

[ampere/watt] loaded by resistor R₁ and amplified by amplification A,then the optical power P_(s) by amplification A, is: $\begin{matrix}{{P_{S}(t)} = \frac{\left\langle {{{\overset{\rightarrow}{E}}_{S}(t)} \cdot {{\overset{\rightarrow}{E}}_{S}(t)}} \right\rangle}{\eta}} & (1)\end{matrix}$The photodetector output current [amperes] is:i(t)=

P _(S)(t)  (2)The photoreceiver output [volts] is thus:v(t)=AR _(L) i(t)=AR _(L) P _(S)(t)  (3)

The output fades only if the optical signal power goes to zero becausethe vector dot product of an optical signal against itself has nopolarization or phase effects. This lack of fading due to polarizationor phase comes at a cost: phase information is lost and signal to noiseratios are severely impacted.

A coherent optical receiver takes advantage of the square lawcharacteristics of photodetectors. A coherent optical receiver combinestwo optical beams, a signal and a local oscillator, together to form aninterference. The interference between these optical waves produces a“beat” which allows the measurement of the phase difference between thesignal and the local oscillator. This interference produces anamplitude, polarization, and phase sensitive receiver output. In orderto consider these effects a discussion of the polarization state ofplane waves is in order. A plane wave contains two orthogonal vectorcomponents which are also orthogonal to the direction of propagation ofthe wave. For purposes of discussion we will consider the plane wave tobe oriented so that the vector components of the electromagnetic fieldlie in an X-Y plane and that the wave propagates in the Z direction.However, this choice of axes is completely arbitrary. In practice, thewave can be oriented in any propagation direction. In order to simplifydiscussions, a simple change of coordinates will make this discussioncompletely general.

The polarization of an electromagnetic (or optical) plane wave, p, isdescribed by a minimum of five parameters. There are two basic ways ofspecifying these parameters. The first way leads to a description whichis oriented towards that which is directly obtained from physicalmeasurements. $\begin{matrix}{{{\overset{\rightarrow}{E}}_{p}\left( {E_{px},E_{py},\Phi_{px},\Phi_{py},\omega_{p},t} \right)} = \begin{bmatrix}{{E_{px}(t)}{\cos\left( {{\omega_{p}t} + \Phi_{px}} \right)}} \\{{E_{py}(t)}{\cos\left( {{\omega_{p}t} + \Phi_{py}} \right)}}\end{bmatrix}} & (4)\end{matrix}$The second manner of describing the polarization state of a wave, p, isoriented more towards the underlying physical mechanisms ofpolarization. See FIG. 1. The description is made in terms of spatialand temporal parameters: $\begin{matrix}{{{\overset{\rightarrow}{E}}_{p}\left( {E_{p},\theta_{p},\psi_{p},\phi_{p},\omega_{p},t} \right)} = {{{{E_{p}(t)}\begin{bmatrix}{\cos\left( \theta_{p} \right)} & {\sin\left( \theta_{p} \right)} \\{- {\sin\left( \theta_{p} \right)}} & {\cos\left( \theta_{p} \right)}\end{bmatrix}}\begin{bmatrix}{\cos\left( \psi_{p} \right)} & 0 \\0 & {\sin\left( \psi_{p} \right)}\end{bmatrix}}\begin{bmatrix}{\cos\left( {{\omega_{p}t} + \phi_{p}} \right)} \\{\sin\left( {{\omega_{p}t} + \phi_{p}} \right)}\end{bmatrix}}} & (5)\end{matrix}$Alternatively, dropping the full variable list in the parentheses andexpanding: $\begin{matrix}{{{\overset{\rightarrow}{E}}_{p}(t)} = \begin{matrix}{{{E_{p}(t)}\begin{bmatrix}{\cos\left( \theta_{p} \right)} & {\sin\left( \theta_{p} \right)} \\{- {\sin\left( \theta_{p} \right)}} & {\cos\left( \theta_{p} \right)}\end{bmatrix}}\begin{bmatrix}{\cos\left( \psi_{p} \right)} & 0 \\0 & {\sin\left( \psi_{p} \right)}\end{bmatrix}} \\{\begin{bmatrix}{\cos\left( \theta_{p} \right)} & {- {\sin\left( \theta_{p} \right)}} \\{\sin\left( \theta_{p} \right)} & {\cos\left( \theta_{p} \right)}\end{bmatrix}\left\lbrack \left. \quad\begin{matrix}{\cos\left( {\omega_{p}t} \right)} \\{\sin\left( {\omega_{p}t} \right)}\end{matrix} \right\rbrack \right.}\end{matrix}} & (6)\end{matrix}$

If E_(p) is constant, the electrical power of this wave can be shown tobe constant and equal to: $\begin{matrix}{{P_{p}(t)} = {\frac{\left\langle {{{\overset{\rightarrow}{E}}_{p}(t)} \cdot {{\overset{\rightarrow}{E}}_{p}(t)}} \right\rangle}{\eta} = \frac{E_{p}^{2}}{2\eta}}} & (7)\end{matrix}$When two waves, S (signal) and L (local oscillator), interfere at theinput of a photoreceiver, the output is: $\begin{matrix}{\begin{matrix}{{v_{out}(t)} = {{AR}_{L}{i(t)}}} \\{= {{AR}_{L}\frac{\left\langle {{{{\overset{\rightarrow}{E}}_{S}(t)} \cdot {{\overset{\rightarrow}{E}}_{S}(t)}} + {{{\overset{\rightarrow}{E}}_{L}(t)} \cdot {{\overset{\rightarrow}{E}}_{L}(t)}} + {2{{{\overset{\rightarrow}{E}}_{L}(t)} \cdot {{\overset{\rightarrow}{E}}_{S}(t)}}}} \right\rangle}{\eta}}}\end{matrix}\begin{matrix}{{v_{out}(t)} = {{v_{L}(t)} + {v_{S}(t)} + {v_{LS}(t)}}} \\{= {{AR}_{L}\left( {{P_{L}(t)} + {P_{S}(t)} + {P_{LS}(t)}} \right)}}\end{matrix}} & (8)\end{matrix}$If the optical power of the local oscillator and signal lightwavesremain constant, a constant photocurrent develops for theself-interference terms (P_(S) and P_(L)) However, if either the localoscillator or the signal lightwaves have any temporal variation inpolarization or phase, the cross interference term (P_(LS)) will be timedependent even if the power of each lightwave remains constant. Solvingfor the cross interference term, we obtain: $\begin{matrix}{{{v_{LS}(t)} = {\frac{{AR}_{L}}{\eta}{E_{L}(t)}{{E_{S}(t)}\begin{bmatrix}{{{\cos({\Delta\theta})}{\cos({\Delta\psi})}{\cos\left( {{{\Delta\omega}\quad t} + {\Delta\phi}} \right)}} +} \\{{\sin({\Delta\theta})}{\sin\left( {2\overset{\_}{\psi}} \right)}{\sin\left( {{{\Delta\omega}\quad t} + {\Delta\phi}} \right)}}\end{bmatrix}}}}{{v_{LS}(t)} = {2{AR}_{L}{\sqrt{{P_{L}(t)}{P_{S}(t)}}\begin{bmatrix}{{{\cos({\Delta\theta})}{\cos({\Delta\psi})}{\cos\left( {{{\Delta\omega}\quad t} + {\Delta\phi}} \right)}} +} \\{{\sin({\Delta\theta})}{\sin\left( {2\overset{\_}{\psi}} \right)}{\sin\left( {{{\Delta\omega}\quad t} + {\Delta\phi}} \right)}}\end{bmatrix}}}}} & (9)\end{matrix}$Where the following definitions are made:Δθ=θ_(S)−θ_(L)Δψ=ψ_(S)−ψ_(L)2 ψ=ψ_(S)+ψ_(L)Δω=ω_(S)−ω_(L)Δφ=φ_(S)−φ_(L)  (10)

The optical cross-interference portion of the receiver output will fadedue to polarization even if the local oscillator and the signallightwaves both do not have zero optical powers. This condition willoccur if:O=cos(Δθ)cos(Δψ)cos(Δωt+Δφ)=sin(Δθ)sin(2 ψ)sin(Δωt+ΔΦ)  (11)Also, equivalently when the condition will occur: $\begin{matrix}{\begin{bmatrix}0 \\0\end{bmatrix} = \begin{bmatrix}{{\cos({\Delta\theta})}{\cos({\Delta\psi})}{\cos\left( {{{\Delta\omega}\quad t} + {\Delta\phi}} \right)}} \\{{\sin({\Delta\theta})}{\sin\left( {2\overset{\_}{\psi}} \right)}{\sin\left( {{{\Delta\omega}\quad t} + {\Delta\phi}} \right)}}\end{bmatrix}} & (12)\end{matrix}$

When heterodyne optical detection is employed (Δω is non-zero, the localoscillator has a different frequency from the signal), the conditionsfor a fade are shown in Table 1. When homodyne detection is employed (Δωis zero), both phase and polarization fading occur. The conditions for ahomodyne fade are shown in Table 2. Heterodyne detection is thereforeseen to be superior to homodyne because the probability of a fade isfully one half as likely. TABLE 1 Heterodyne Fading Conditions Type ofFade Required Simultaneous (k is an integer Conditions for a Fade toOccur Orthogonal Rotation and Δ

= (2k + 1)π/2 ψ_(S) + ψ_(L) = 0 Opposite Ellipticity Orthogonal Rotationand Equal Δ

= (2k + 1)π/2 ψ_(S) + ψ_(L) ± π Circular Ellipticity Equal Rotation andOrthogonal Δ

= 0 Δψ = ±π/2 Ellipticity Opposite Rotation and Δ

= ±π Δψ = ±π/2 Orthogonal Ellipticity

TABLE 2 Homodyne Fading Conditions Type of Fade Required Simultaneous (kand m are integers) Conditions for a Fade to Occur Orthogonal Rotationand Δ

= (2k + 1)π/2 ψ_(S) + ψ_(L) = 0 Opposite Ellipticity Orthogonal Rotationand Equal Δ

= (2k + 1)π/2 ψ_(S) + ψ_(L) ± π Circular Ellipticity Equal or OppositeRotation and Δ

= kπ Δψ = ±π/2 Orthogonal Ellipticity Orthogonal Rotational and Equal Δ

= (2k + 1)π/2 Δφ = mπ or Opposite Phase

Given the conditions for and the functional relation of a fade, thequestion now arises as to how a fade can be prevented. Since the signalis being measured, no a priori knowledge is assumed and therefore E_(S),θ_(S), Ψ_(S), Φ_(S) are all probably unknown quantities. If fading isprevented, then no loss of information occurs and determination of thesefour parameters is possible. In order to decode the optical receiveroutput into these parameters, at least four independent measurementsmust be made to uniquely determine these four independent variables.However, if the interfering optical beam (or beams) of the localoscillator are unknown, then additional independent measurements must bemade (four additional measurements for each unknown beam) to determinethe E_(L), θ_(L), Ψ_(L), or Φ_(L) for each optical beam of the localoscillator. The cross-reference output of the photoreceiver, v_(LS) (t),offers the only means by which to measure these parameters. If theparameters cannot be determined from this output, then an optical fadecannot be ruled out.

We shall now examine the information which can be gleaned from thisoutput. Define the following functions. $\begin{matrix}{\begin{matrix}{{v_{l}\left( {E_{L},E_{S},{\Delta\theta},{\Delta\psi}} \right)} = {\frac{{AR}_{L}}{2\eta}{E_{L}(t)}{E_{S}(t)}{\cos({\Delta\theta})}{\cos({\Delta\psi})}}} \\{= {{AR}_{L}\sqrt{{P_{L}(t)}{P_{S}(t)}}\quad{\cos({\Delta\theta})}{\cos({\Delta\psi})}}}\end{matrix}\begin{matrix}{{v_{Q}\left( {E_{L},E_{S},{\Delta\theta},{2\overset{\_}{\psi}}} \right)} = {\frac{{AR}_{L}}{2\eta}{E_{L}(t)}{E_{S}(t)}{\sin({\Delta\theta})}{\sin\left( {2\overset{\_}{\psi}} \right)}}} \\{= {{AR}_{L}\sqrt{{P_{L}(t)}{P_{S}(t)}}\quad\sin({\Delta\theta}){\sin\left( {2\overset{\_}{\psi}} \right)}}}\end{matrix}} & (13)\end{matrix}$In the homodyne case (Δω is zero), we obtain the following output:v _(LS)(t)=2AR _(L)

√{square root over (P _(L)(t)P_(S)(t))}(cos(Δθ)cos(Δψ)cos(Δφ)+sin(Δθ)sin(2 ψ)sin(Δφ))v _(LS)(t)=2v_(l)(E _(L) ,E _(S),Δθ,Δψ)cos(Δφ)+2v _(Q)(E _(L) ,E _(S),Δθ,2ψ)sin(Δφ)  (14)The homodyne output only allows the measurement of one quantity. Theoutput provides only one independent measurement (one equation) whereasa minimum of four are required. In the heterodyne case (Δω is non-zero),the output is: $\begin{matrix}{{{v_{LS}(t)} = {2{AR}_{L}\sqrt{{P_{L}(t)}{P_{S}(t)}}\begin{pmatrix}{{{\cos({\Delta\theta})}{\cos({\Delta\psi})}{\cos\left( {{{\Delta\omega}\quad t} + {\Delta\phi}} \right)}} +} \\{{\sin({\Delta\theta})}{\sin\left( {2\overset{\_}{\psi}} \right)}{\sin\left( {{{\Delta\omega}\quad t} + {\Delta\phi}} \right)}}\end{pmatrix}}}{{v_{LS}(t)} = {\frac{{AR}_{L}}{2\eta}{E_{L}(t)}{E_{S}(t)}\begin{pmatrix}{{{\cos({\Delta\theta})}{\cos({\Delta\psi})}{\cos\left( {{{\Delta\omega}\quad t} + {\Delta\phi}} \right)}} +} \\{{\sin({\Delta\theta})}{\sin\left( {2\overset{\_}{\psi}} \right)}{\sin\left( {{{\Delta\omega}\quad t} + {\Delta\phi}} \right)}}\end{pmatrix}}}{{v_{LS}(t)} = {{2{v_{1}\left( {E_{L},E_{S},{\Delta\theta},{\Delta\psi}} \right)}{\cos\left( {{{\Delta\omega}\quad t} + {\Delta\phi}} \right)}} + {2{v_{Q}\left( {E_{L},E_{S},{\Delta\theta},{2\quad\overset{\_}{\psi}}} \right)}{\sin\left( {{{\Delta\omega}\quad t} + {\Delta\phi}} \right)}}}}} & (15)\end{matrix}$

Since sine and cosine waves are orthogonal, the heterodyne receiverprovides two independent measurements by mixing down to baseband the Δωradian frequency components. Thus, two outputs are obtained:V _(l)(t)=<v _(LS)(t)cos(Δωt)>=v _(l)(E _(L)(t),E_(S)(t),Δθ(t),Δψ(t))cos(Δφ(t))V _(Q)(t)=<v _(LS)(t)sin(Δωt)>=v _(Q)(E _(L)(t),E _(S)(t),Δθ(t),2ψ(t))sin(Δφ(t))  (16)

b. Correlation or Time-Delay Multiplexing

In many optical sensor applications, the lightwave signalheterodyne-detected by the photodetector system is a composite opticalsignal formed from the superposition of many individual optical signals.When the receiver lightwave is generated by backscatter, the compositeoptical signal is the superposition of individual light signalsgenerated by a continuum of reflections of an interrogation lightsource. The temporal and spatial characteristics of each reflector orreflective region creates a modulation of the interrogation lightsource. The time-delay, amplitude, polarization and phase states controlthe backscattered-modulation of these individual optical signalsarriving at the photodetector with a unique time-delay interval can beseparated into channels which sort the optical signals into time-delayslots or bins. Depending upon how the signals are generated, thesechannels can represent spatial regions in space or time-delay slots of atime-domain reflectometer mechanism.

Let an interrogation lightwave source be generated by modulating theamplitude, phase or polarization of a coherent lightwave with atime-structured correlation code, c(t). The correlation code, c(t) canbe a series of pulses, chirps, binary sequences or any other type ofcode which provides the required correlation characteristics. If thelightwave source is:E _(SS)(t)=E _(SS) cos(ω_(S) t)  (17)Then an amplitude modulated interrogation source is:E _(i)(t)=μ_(A) c(t)E _(ss) cos(ω_(s) t)  (18)Alternatively, a phase modulated interrogation source is:E _(i)(t)=E _(SS) cos(ω_(S) t+μ_(p) c(t)).  (19)If c(t) is chosen to be temporally structured properly, then:$\begin{matrix}{{R_{i}(\tau)} = {\left\langle {{E_{i}(t)}{E_{i}\left( {t + \tau} \right)}} \right\rangle \approx \left\{ \begin{matrix}{\frac{E_{SS}^{2}}{2};} & {\tau \approx 0} \\{0;} & {otherwise}\end{matrix} \right.}} & (20)\end{matrix}$c(t) must be chosen so that an a priori decoding/demultiplexingfunction, d(t), exists such that: $\begin{matrix}{{b\left( {t,\tau} \right)} = {\left\langle {{d(t)}{E_{i}\left( {t + \tau} \right)}} \right\rangle \approx \left\{ \begin{matrix}{{\xi\quad E_{SS}{\cos\left( {{{\Delta\omega}\quad t} + \phi} \right)}};} & {\tau \approx 0} \\{0;} & {otherwise}\end{matrix} \right.}} & (21)\end{matrix}$

For instance, suppose the interrogation wave is: $\begin{matrix}{{{E_{i}(t)} = {\mu_{A}{c(t)}E_{ss}{\cos\left( {\omega_{S}t} \right)}}}{{and}\text{:}}} & (22) \\{{R_{c}(\tau)} = {\left\langle {{c(t)}{c\left( {t - \tau} \right)}} \right\rangle \approx \left\{ \begin{matrix}{1;{\tau \approx 0}} \\{0;{\tau \neq 0}}\end{matrix} \right.}} & (23)\end{matrix}$then a valid decoding and temporal and spatial domains demultiplexingfunction is: $\begin{matrix}\begin{matrix}{{d(t)} = {\mu_{d}{C(t)}E_{L}{\cos\left( {{\left( {{\Delta\omega} + \omega_{S}} \right)t} + \phi} \right)}}} \\{{b\left( {t,\tau} \right)} = \left\langle {{d\left( {t - \tau} \right)}{E_{i}(t)}} \right\rangle} \\{= \left\{ \begin{matrix}{{\frac{\mu_{d}\mu_{A}E_{SS}E_{L}}{2}\cos\quad\left( {{{\Delta\omega}\left( {t - \tau} \right)} + \phi - {\omega_{S}\tau}} \right)};} & {\tau = 0} \\{0;} & {otherwise}\end{matrix} \right.}\end{matrix} & (24)\end{matrix}$

Therefore, delaying the correlation decoding/demultiplexing functiond(t) allows demultiplexing of delay multiplexed signals identifiable byspeed of propagation and distance of flyback travel. Suppose an opticalwave is formed a summation of delayed signals modulated onto theinterrogation wave E_(i)(t), then the received wave, E_(b)(t), is:$\begin{matrix}{{E_{b}(t)} = {\sum\limits_{n = 1}^{N}{{A_{n}\left( {t - \tau_{n}} \right)}\mu_{A}{c\left( {t - \tau_{n}} \right)}E_{SS}{\cos\left( {{\omega_{S}\left( {t - \tau_{n}} \right)} + {\Phi_{n}\left( {t - \tau_{n}} \right)}} \right)}}}} & (25)\end{matrix}$

Then multiplying by the decoding/demultiplexing function, d(t−τ_(m)), weobtain: $\begin{matrix}\begin{matrix}{{d(t)} = {\mu_{d}{c(t)}E_{L}{\cos\left( {{\left( {{\Delta\quad\omega} + \omega_{S}} \right)t} + \phi} \right)}}} \\{{b\left( {t,\tau_{m}} \right)} = \left\langle {{d\left( {t - \tau_{m}} \right)}{E_{b}(t)}} \right\rangle} \\{{b\left( {t,\tau_{m}} \right)} \approx {\frac{\mu_{d}\mu_{A}E_{SS}E_{L}}{2}{A_{m}\left( {t - \tau_{m}} \right)}}} \\{{\cos\left( {{{\Delta\omega}\left( {t - \tau_{m}} \right)} + \phi - {\omega_{S}\tau_{m}} + {\Phi_{m}\left( {t - \tau_{m}} \right)}} \right)}.}\end{matrix} & (26)\end{matrix}$Because τ_(n) is unique, the amplitude signal A_(m)(t−τ_(m)) and thephase signal Φ_(m)(t−τ_(m)) are both extracted from E_(b)(t) bymultiplying by the decoding/demultiplexing function, d(t−τ_(m)). Thetechnique is applicable to a wide variety of other optical signalmultiplexing applications. Specifically, the technique can be used tospatially separate optical signals arriving from a temporally varyingtime-domain reflectometer optical backscatter process from an array offiber optic acoustic sensors.(2) Description and Operation of the Rayleigh Optical Scattering andEncoding (ROSE) System

a. ROSE Optical Phase Sensor Interrogation Enables Sensor Subsystem

In order to more fully describe the capabilities and new features of theinvention, the application of the invention to a subsystem 1, FIG. 2, ofROSE which launches an interrogation signal onto fiber span 9 andretrieves lightwave back propagation from a continuum of locations alongthe span. Back propagation mechanisms may include Rayleigh OpticalScattering (ROS) and other effects generated within the optical fiber.Rayleigh Optical Scattering (ROS) in an optical fiber backscatters lightincident upon the fiber. The incident light transverses down the opticalfiber to the scattering point/region. At the scattering region theincident light is backscattered back up the optical fiber. As the lighttransverses the round trip optical path (i.e., distance of flybacktravel) any disturbance of the fiber which increase or decrease theoptical path length will cause the phase of the incident andbackscattered light to be modulated. Suppose a pressure is applied tothe optical fiber. The pressure elongates the path length of the lighttransversing the region.

Refer to FIG. 2. for the following discussion. In the FIGS. like partscorrespond to like numbers. Let p(t,z) be pressure applied to theoutside of the optical fiber at time, t, and at point or length, z,along the fiber axis. Then if an interrogation optical wave, E_(i)(t),generated by laser 3, passed through optical coupler 4 and modulated byoptical modulator 5 is applied to optical coupler 7, this results in thefollowing output interrogation wave, E_(i)(t), being transmitted downthe fiber 9:E _(i)(t)=μ_(A) c(t)E _(ss) cos(ω_(S) t)  (27)

The backscattered wave, E_(b)(t), arriving back at an optical coupler 7from ROSE fiber optic array 9 passes into optical path 11. Thebackscattered light which arrives at optical path 11 is the summation ofall light backscattered from a continuum of locations along the lengthof the ROSE fiber optic span 9. As will later herein be described indetail, fiber 9 has a longitudinal strain component enhancing coating12. If r(z) is the reflection density at point or length z along thefiber and c_(L) is the optical wave speed, then the backscattered lightafter a pressure p(t,z) is applied to fiber is representedmathematically as: $\begin{matrix}{{{E_{b}(t)} = {\int_{0}^{\infty}{{r\left( {\hat{z}\left( {t,z} \right)} \right)}\mu_{A}{c\left( {t - \frac{2{\hat{z}\left( {t,z} \right)}}{c_{L}}} \right)}E_{SS}{\cos\left( {\omega_{S}\left( {t - \frac{2{\hat{z}\left( {t,z} \right)}}{c_{L}}} \right)} \right)}{\mathbb{d}z}}}}{{where}\text{:}}} & (28) \\{{\hat{z}\left( {t,z} \right)} = {z + {\mu_{L}{\int_{0}^{z}{{p\left( {t,x} \right)}{{\mathbb{d}x}.}}}}}} & (29)\end{matrix}$

If the distributed reflection, r(z) is essentially independent of theapplied pressure, p(t,z) then the backscatter is: $\begin{matrix}{{E_{b}(t)} = {\int_{0}^{\infty}{{r(z)}\mu_{A}{c\left( {t - \frac{2{\hat{z}\left( {t,z} \right)}}{c_{L}}} \right)}E_{SS}{\cos\left( {\omega_{S}\left( {t - \frac{2{\hat{z}\left( {t,z} \right)}}{c_{L}}} \right)} \right)}{{\mathbb{d}z}.}}}} & (30)\end{matrix}$

Since optical path length change caused by the applied pressure, p(t,z)is usually extremely small (on the order of 10⁻⁶ to 10¹ times an opticalwavelength), the backscattered light from each z distance down the fiberarrives at the optical path 11 with a transversal delay, τ(t,z), equalto: $\begin{matrix}{{\tau\left( {t,z} \right)} \approx {\frac{2z}{c_{L}\quad}.}} & (31)\end{matrix}$

Therefore, to receive the signal S₁ backscattered from the fiber regionat length-down-the-fiber z=L₁, the correlational multiplexingcharacteristic of the transmitted interrogation light can be utilized.Multiplication of the total backscattered optical signal by thecorrelation decoding/demultiplexing function, d(t−τ(t,z₁)), produces anoutput which contains the signal, S₁, backscattered from a distance L₁down the fiber and rejects signals originating from other fiber regions,such as S₂, S_(n) and etc. Representing this process mathematically, theresulting channel output, B(t, L₁) is obtained as follows:$\begin{matrix}\begin{matrix}{{b\left( {t,\tau_{1}} \right)} = \left\langle {{d\left( {t - \tau_{1}} \right)}{E_{b}(t)}} \right\rangle} \\{= \left\langle {{d\left( {t - \frac{2L_{1}}{c_{L}}} \right)}{E_{b}(t)}} \right\rangle} \\{= {B\left( {t,L_{1}} \right)}} \\{{d\left( {t - \frac{2L_{1}}{c_{L}}} \right)} = {\mu_{d}{c\left( {t - \frac{2L_{1}}{c_{L}}} \right)}E_{L}{\cos\left( {{\left( {{\Delta\omega} + \omega_{S}} \right)\left( {t - \frac{2L_{1}}{c_{L}}} \right)} + \phi} \right)}}} \\{{E_{b}(t)} = {\int_{0}^{\infty}{{r(z)}\mu_{A}{c\left( {t - \frac{2z}{c_{L}}} \right)}E_{SS}{\cos\left( {\omega_{S}\left( {t - \frac{2{\hat{z}\left( {t,z} \right)}}{c_{L}}} \right)} \right)}{\mathbb{d}z}}}} \\{{\Phi\left( {z,L_{1}} \right)} = {\phi - \frac{2\left( {{\Delta\omega} + \omega_{S}} \right)L_{1}}{c_{L}} + {{\Delta\omega}\quad\frac{2z}{c_{L}}}}}\end{matrix} & (32) \\\begin{matrix}{{B\left( {t,L_{1}} \right)} = {\mu_{d}\mu_{A}E_{L}E_{SS}{\int_{0}^{\infty}{{r(z)}{R_{c}\left( \frac{2\left( {z - L_{1}} \right)}{c_{L}} \right)}{\cos\left( {{{\Delta\omega}\quad t} + {\Phi\left( {z,L_{1}} \right)} + {\frac{2\mu_{L}\omega_{S}}{c_{L}}{\int_{0}^{z}{{p\left( {t,x} \right)}{\mathbb{d}x}}}}} \right)}z{\mathbb{d}z}}}}} \\{{\Delta\quad{\Phi\left( {t,z} \right)}} = {{\Phi\left( {z,L_{1}} \right)} + {\frac{2\mu_{L}\omega_{S}}{C_{L}} \cdot {\int_{0}^{z}{{p\left( {t,x} \right)}{\mathbb{d}x}}}}}} \\{{B\left( {t,L_{1}} \right)} = {v_{E}{\int_{0}^{\infty}{{r(z)}{R_{c}\left( \frac{2\left( {z - L_{1}} \right)}{c_{L}} \right)}{\cos\left( {{{\Delta\omega}\quad t} + {{\Delta\Phi}\left( {t,z} \right)}} \right)}{\mathbb{d}z}}}}} \\{{B\left( {t,L_{1}} \right)} = {V_{E}r_{L_{1}}{\cos\left( {{{\Delta\omega}\quad t} + \Phi_{L_{1}} + {\frac{2\mu_{L}\omega_{S}}{c_{L}} \cdot {\int_{0}^{L_{1}}{{p\left( {t,x} \right)}{\mathbb{d}x}}}}} \right)}}}\end{matrix} & (33)\end{matrix}$Because of the correlation properties of the interrogation light, theautocorrelation function R_(c)(τ) is very small at all spatial locationsexcept those in the vicinity of z=L₁. Therefore, all signals originatinganywhere else are rejected. Furthermore, the phase of the channel outputat location L₁ will be the summation or integration of all pressurechanges along the bi-directional transversal path. This unusualphenomenon has been demonstrated with experimental hardware.

Once the correlation process isolates the optical signal originatingfrom a spatial region, the signal must be phase demodulated to extractthe pressure information. The signal is I (in phase) and Q (quadraturephase) demodulated is: $\begin{matrix}\begin{matrix}{{B_{1}\left( {t,L_{1}} \right)} = \left\langle {{B\left( {t,L_{1}} \right)}{\cos\left( {\Delta\quad\omega\quad t} \right)}} \right\rangle} \\{{B_{1}\left( {t,L_{1}} \right)} \approx {V_{E}r_{L_{1}}{\cos\left( {\Phi_{L_{1}} + {\frac{2\mu_{L}\omega_{S}}{c_{L}} \cdot {\int_{0}^{L_{1}}{{p\left( {t,x} \right)}{\mathbb{d}x}}}}} \right)}}} \\{= {V_{1}{\cos\left( \Phi_{1} \right)}}} \\{{B_{Q}\left( {t,L_{1}} \right)} = \left\langle {{B\left( {t,L_{1}} \right)}{\sin\left( {\Delta\quad\omega\quad t} \right)}} \right\rangle} \\{{B_{Q}\left( {t,L_{1}} \right)} \approx {{- V_{E}}r_{1}{\sin\left( {\Phi_{L_{1}} + {\frac{2\mu_{L}\omega_{S}}{c_{L}} \cdot {\int_{0}^{L_{1}}{{p\left( {t,x} \right)}{\mathbb{d}x}}}}} \right)}}} \\{= {{- V_{1}}{{\sin\left( \Phi_{1} \right)}.}}}\end{matrix} & (34)\end{matrix}$

Then I & Q, or cosine phase and sine phase outputs are converted intoeither phase rate or phase outputs with simple analog or digitalhardware. The phase, so demodulated, allows the inference of informationabout the acoustic pressure down the fiber to the measurement point.

Once the I & Q outputs are generated, the temporal phase state of B(t,L₁) can be determined by one of several types of phase demodulationprocesses. The phase state of the region of L₁ spatial delay istherefore: $\begin{matrix}{\Phi_{1} = {\Phi_{L_{1}} + {\frac{2\mu_{L}\omega_{S}}{c_{L}} \cdot {\int_{0}^{L_{1}}{{P\left( {t,x} \right)}{{\mathbb{d}x}.}}}}}} & (35)\end{matrix}$

Likewise, the plurality (which may be a large number, e.g., 5000) ofoptical signals arising with spatial delays, such as the propagationtime for flyback travel to L₂ or L_(n), can be correlated out of thebackscattered signal E_(b)(t). These are: $\begin{matrix}\begin{matrix}{{B\left( {t,L_{2}} \right)} \approx {V_{E}r_{L_{2}}{\cos\left( {{\Delta\quad\omega\quad t} + \Phi_{L_{2}} + {\frac{2\mu_{L}\omega_{S}}{C_{L}} \cdot {\int_{0}^{L_{2}}{{p\left( {t,x} \right)}{\mathbb{d}x}}}}} \right)}}} \\{{B\left( {t,L_{n}} \right)} \approx {V_{E}r_{L_{n}}{\cos\left( {{\Delta\quad\omega\quad t} + \Phi_{L_{n}} + {\frac{2\mu_{L}\omega_{S}}{C_{L}} \cdot {\int_{0}^{L_{2}}{{p\left( {t,x} \right)}{\mathbb{d}x}}}}} \right)}}}\end{matrix} & (36)\end{matrix}$With corresponding phase signals of: $\begin{matrix}\begin{matrix}{\Phi_{2} = {\Phi_{L_{2}} + {\frac{2\mu_{L}\omega_{S}}{c_{L}} \cdot {\int_{0}^{L_{2}}{{p\left( {t,x} \right)}{\mathbb{d}x}}}}}} \\{\Phi_{n} = {\Phi_{L_{n}} + {\frac{2\mu_{L}\omega_{S}}{c_{L}} \cdot {\int_{0}^{L_{n}}{{p\left( {t,x} \right)}{{\mathbb{d}x}.}}}}}}\end{matrix} & (37)\end{matrix}$

The phase signals, obtained by phase demodulation of each B(t,L_(m)),represent a pressure field p(t,z) which is integrated along the length,z, of the fiber. Therefore, rather than directly measure p(t,z) thesensor provides all of the accumulated pressure effects down the fiberto the measurement point, L_(m) (where m is integer corresponding to themeasurement point). In sensor arrays, it is usually desired to detectthe pressure over a specific measurement region. If two optical signalsS_(j) and S_(k) are received from measurement lengths L_(j) and L_(k),the corresponding demodulated phases Φ_(j) and Φ_(k) are:$\begin{matrix}\begin{matrix}{\Phi_{j} = {\Phi_{L_{j}} + {\frac{2\mu_{L}\omega_{S}}{c_{L}} \cdot {\int_{0}^{L_{j}}{{p\left( {t,x} \right)}{\mathbb{d}x}}}}}} \\{\Phi_{k} = {\Phi_{L_{k}} + {\frac{2\mu_{L}\omega_{S}}{c_{L}} \cdot {\int_{0}^{L_{k}}{{p\left( {t,x} \right)}{{\mathbb{d}x}.}}}}}}\end{matrix} & (38)\end{matrix}$

A sensor between the lengths down the fiber of L_(j) and L_(k)(L_(k)>L_(j)) is formed by subtracting the two phases: $\begin{matrix}\begin{matrix}{{\Phi_{k} - \Phi_{j}} = {{\Delta\quad\Phi_{kj}} = {\left( {\Phi_{L_{k}} + {\frac{2\mu_{L}\omega_{S}}{c_{L}} \cdot {\int_{0}^{L_{k}}{{p\left( {t,x} \right)}{\mathbb{d}x}}}}} \right) -}}} \\{\left( {\Phi_{L_{1}} + {\frac{2\mu_{L}\omega_{S}}{c_{L}} \cdot {\int_{0}^{L_{j}}{{p\left( {t,x} \right)}{\mathbb{d}x}}}}} \right)} \\{{\Delta\quad\Phi_{kj}} = {\Phi_{L_{k}} - \Phi_{L_{j}} + {\frac{2\mu_{L}\omega_{S}}{c_{L}} \cdot \left( {{\int_{0}^{L_{k}}{{p\left( {t,x} \right)}{\mathbb{d}x}}} - {\int_{0}^{L_{j}}{{p\left( {t,x} \right)}{\mathbb{d}x}}}} \right)}}} \\{{\Delta\Phi}_{kj} = {\Phi_{L_{k}} - \Phi_{L_{j}} + {\frac{2\mu_{L}\omega_{S}}{c_{L}} \cdot {\int_{L_{j}}^{L_{k}}{{p\left( {t,x} \right)}{\mathbb{d}x}}}}}} \\{{\Delta\Phi}_{kj} = {{\Delta\Phi}_{L_{k}L_{j}} + {\frac{2\mu_{L}\omega_{S}}{c_{L}} \cdot {\int_{L_{j}}^{L_{k}}{{p\left( {t,x} \right)}{{\mathbb{d}x}.}}}}}}\end{matrix} & (39)\end{matrix}$

The resulting sensor is of length ΔL=L_(k)−L_(j) with a center positionof (L_(k)+L_(j))/2. The differencing of phase signals Φ_(j) and Φ_(k)into a new phase signal ΔΦ_(kj), allows a virtual sensor of arbitraryposition and length to be formed. The resulting spatially differentialsensor also adds the advantage of minimizing other effects such aslead-in fiber strum or vibration which create unwanted phase signals.

The above phenomena illustrates that when the interrogation light isproperly encoded, a ROSE (Rayleigh Optical Scattering and Encoding)sensor system is enabled. The subject invention therefore enables theROSE concept. The subject invention enables spatial discrimination ofthe optical backscatter effects in a ROSE sensor. The spatialdifferencing technique rejects unwanted common mode signalsinadvertently introduced in fiber leads down to the sensor region. Theinvention also applies in a similar manner to more conventional fiberoptic acoustic sensor arrays (i.e., those having Bragg reflectivegrating sensors) or to non-fiber optic remote optical sensors whichdetect phase.

b. Pointwise Signal Delay Multiplexing

The invention also applies to point-wise non-distributed sensors orartificially generated multiplexing by electronics means. Theinterrogation lightwave can be intercepted and retransmitted back to thereceiver with an artificial, electronically generated delay, as a meansof delay/correlation multiplexing many channels.

(3) Description of a Fiber System Implementation

The invention can be realized with bulk optical, fiber optical orintegrated optical components. For simplicity, a fiber opticimplementation will be presented. However, the fiber optic embodiment isbeing presented without intent of limitation. The teachings of theinvention can be used to implement a reflectometer system in accordancewith the present invention using these and other instrumentalitiesproviding a light path that has the innate property of producing backpropagation of portions of an interrogation signal at a continuum oflocations along the length of the propagation path therethrough.

a. Optical Transmitter and Time-Delay Multiplexing Process.

FIG. 3 is an illustrative block diagram implemention of the Rayleighoptical scattering and endcoding (ROSE) sensor system 2. Like partscorrespond to like numbers. A lightwave from transmitter laser, 3, ispropagated through optical coupler or beamsplitter, 4. The smallerportion of the transmitter laser power split off by optical coupler, 4,is passed by optical path, 39, to the phase locking means opticalreceiver 35. The larger portion of the transmitter laser light power issplit by optical coupler, 4, and propagated to optical modulator, 5. Theoptical modulator, 5, modulates the laser light passing from opticalcoupler, 4, with correlation code, c(t), as electronically generated inmaster correlation code generator, 53, and amplified by amplifier, 49.The correlation code, c(t), is modulated onto the laser light in opticalmodulator, 5. This modulated light comprises the optical interrogationlightwave, E_(i)(t). The optical modulator, 5, may modulate theamplitude, polarization or phase of the laser light subject to theteachings of the invention. The interrogation lightwave is propagatedfrom optical modulator, 5, to optical coupler, beamsplitter orcirculator, 7. The interrogation lightwave passes through the opticalcoupler, 7, into optical fiber or other light propagation medium, 9.Hereinafter, “down”, indicates a transversal on the optical path, 9,away from coupler, 7; “up” indicates a transversal on the optical path 9toward the optical coupler, beamsplitter or circulator, 7. Theinterrogation lightwave which transverses down the optical fiber ormedium 9 is modulated and is backscattered or returned by other meanswith equivalent optical path lengths (equivalent to a time delay), L₁,L₂ . . . L_(n) corresponding to sensors or multiplexed channels S₁, S₂ .. . S_(n). The returned interrogation lightwave is a composite opticalsignal modulated by signals due to the S₁ through S_(n) modulating andtime-domain multiplexing actions.

More particularly, the propagation of the optical spread-spectruminterrogation signal down the continuous full span of the optical fiberspan, signal launch end to remote end, causes a back-propagatingcomposite optical signal, which is the linear summation, or integrationspatially, of all of the individual, continuous, or continuum ofback-reflections along the span of the optical fiber.

One component of this composite signal is comprised of the naturallyoccurring continuum of optical back reflections (including Rayleighoptical scattering ((ROS)) effects) of the optical spread spectrumcarrier signal that is formed by modulating the primary carrier signalby the spectrum spreading signals. Another component is comprised of theartificially occurring optical back reflections, either-point wisereflections or distributed reflections, of the optical spread spectrumcarrier signal that is formed due to propagation discontinuities as theresult of presence of a fiber cable coupler in span 9. Still anothercomponent comprised of the continuum of modulations at locations alongthe span of the reflected signals due to longitudinal components ofoptical path length change, causing a delay in the reflected signal,experienced by the fiber optical span along its length.

Such optical path length change or delay may be caused by a variety ofpossible sources including acoustic pressure waves incident to thefiber, electromagnetic fields coupled to the fiber, mechanical strain orpressure on the fiber, thermal strain or pressure induced in the fiber,or other means of causing change in the optical path length. Use of theacoustic pressure waves mode of changing path length in perimeterintrusion monitoring systems is the principle embodiment illustratedherein. In this use, optical fiber span 9 is employed to provide anarray of virtual geophones buried at a range of depths beneath thesurface of the ground of about between six (6) inches and one (1) foot,to sense motion of an object on the surface of the ground. The acousticpressure wave sensing mode is also useful to sense seismic signals, asfor example as linear arrays inserted into casing structures of anexisting oil wells. Predetermined artificial pressure wave producingshocks are imparted into the ground, and the responses from the sensorare used to locate secondary oil deposits. The acoustic pressure wavesensing mode is further useful for employing span 9 as an array ofvirtual hydrophones, with the media which couples the signals to thehydrophones at least in part being the body of water in which the arrayis immersed. Such hydrophone arrays find use as naval undersea warfaretowed arrays, or towed geophysical exploration arrays. In the latter thearrays respond to artificially produced shocks of predeterminedcharacter and location induced in the body of water, and the response ofthe array to bottom return signals are used to locate ocean bottomgeophysical feature indicating likely presence of an oil deposit. Yetfurther, a sensing position on a fiber span 9 could be used to receiveas an input microphonic signals suitably imparted to the region of thesensing position. The electromagnetic field sensing mode of fiber span 9could be used for monitoring electronic signals along atelecommunication cable's span to localize malfunctions. Responses offiber span 9 to mechanical, pressure or thermal strains can be used insystems for monitoring such strains.

The composite lightwave propagates up the optical fiber or medium 9,passes through optical coupler, beamsplitter or circulator, 7, tooptical pathway, 11. Optical pathway, 11, passes the backscattered,time-delay multiplexed, composite lightwave, E_(b)(t), to the opticalreceiver, 15.

Preferably, fiber 9 is of the relatively low cost, conventionalsingle-mode or multimode, fiber cable types.

Further it is preferable that fiber 9 have extruded thereon a coating 12of a material which enhances the longitudinal strain which the fiberundergoes from a given radially, or generally laterally, appliedpressure wave strain. Materials which provide such enhancement includeextrudable thermoplastic polymers (TPU's) or extrudable thermoplasticelastomers (TPE's) which exhibit a combination of a low Young's modules(E) and a low Poisson's ratio (a). The Poisson's ration is preferablybelow 0.5, which is the Poisson's ratio of natural rubber. Examples ofsuch materials include: (i) low density polyethylene, havingcharacteristic E=1.31 dynes/cm²×10⁻¹⁰ and σ=0.445; and (ii) polystyrene,E=3.78 dynes/cm² and σ=0.35 (values as reported in the paper, R. Hughesand J. Javzynski, “Static Pressure Sensitivity Amplification inInterferometric Fiber-Optic Hydrophones,” Applied Optical/Vol. 19/No.1/1 January 1980).

An alternate embodiment of fiber 9, albeit involving significantlygreater cost per unit length of the fiber, is to provide fiber in themore expensive form of a polarization preserving or single polarization,optical fiber. The polarization preserving fiber of this type holds thebackscattering light in a narrow range of polarization states so that asubstantially single RF signal 21 enters a single set 23 of correlators,reducing the complexity of the system. However, in cases involving longsurveillance lines this alternative embodiment becomes expensive in costof fiber.

The correlation code generator 53 creates a signal, c(t), that has abroad bandwidth. The broadband nature of the correlation code isrequired to obtain the desired properties in the signals autocorrelationfunction. The calculation and definition of the autocorrelation functionof any general signal is well known and defined in signal processingliterature. The correlation code signal, c(t), is so structured that itsautocorrelation function is highly peaked at zero delay, and is verysmall away from zero delay. This criterion is well known to those ofskill in the art and is the essence of why the correlation code has abroad bandwidth. Any signal that has the desired autocorrelationfunction properties can be used as the correlation code in theinvention. There are many reasons for choosing one correlation code overanother: ease of creation; autocorrelation properties; cost of creationhardware; cost of correlation hardware; and effectiveness in producingspread spectrum signal effects. According to the teaching of theinvention, the correlation code for the invention can be a binarysequence with a desired transorthogonal autocorrelation property(sometimes called a pseudonoise sequence), a pseudorandom number (PRN)sequence with the such desired autocorrelation property, chirps, orother types of signals which provide correlations code having predicablenon-repetitive behavior. The foregoing list of types of sequence signalswhich may be employed to modulate the carrier lightwave signal includesboth “binary pseudonoise sequences” and “pseudorandom number (PRN)sequences.” For purposes of construction of this specification and theappended claims, these terms are employed as they are defined under thelistings “Pseudonoise (PN) sequence (communication satellite)” and“Pseudorandom number sequence” at pages 747 and 748 of the “IEEEStandard Dictionary of Electrical and Electronic Terms” (FourthEdition), which listings are hereby incorporated herein by reference.Further for purposes of construction of this specification and theappended claims, it is deemed that “binary pseudonoise sequence” isgeneric and “pseudorandom number sequence” is a species thereof. Stillfurther for purposes of construction of this specification and itsappended claims, both terms are deemed to include analog signal forms ofsequences as well as digital signal forms.

It is to be appreciated that in addition to its correlation encodingfunction, master correlating code generator 53 is a source of aspectrum-spreading signal comprised of a spectrum-spreading signal whichproduces an autocorrelation that is well behaved. It has one dominatemaxima at zero correlation delay, and its spectrum is large enough toprovide sampling of the said optical fiber spatially along the length ofthe fiber 9 with a resolution commensurate with a sub-length ΔZ of fiberspan 9. These characteristics enable segmentation of an optical fiber 9of span length L into n segments in accordance with a relationshipL<ΔZ·n.  (40)In this relationship ΔZ is a segment length of the fiber span whoselength is one-half the distance traveled by light propagating throughone zero delay temporal time span of the autocorrelation maxima, ΔT,such that C_(L) is the speed of light in the said optical fiber and ΔTis approximately equal to the reciprocal of the spread signal opticalbandwidth.

An illustrative embodiment of generator 53 is a shift register typepseudorandom number code generator, having n bits, wherein a code isgenerated that satisfies said resolution sublength and segment lengthrelationship by choosing an appropriate combination of the number of itsbits and the clock time.

The temporal length of the code sequence which is reiteratively producedby generator 53 may be either less than the time period for propagationof a lightwave to the remote end of span and propagation back of abackscattering (i.e. distance of flyback travel), or greater than thistime period. It cannot be equal to this period.

The predetermined timing base employed by the source of the spectrumspreading signals, which determines the length of ΔZ segment is sochosen to provide a positive enhancement to the ratio of the power ofback propagating Rayleigh scattering effect P_(R) to the power of theforward propagated Rayleigh scattering effect P_(T), in accordance withthe following equation: $\begin{matrix}{{\frac{P_{r}}{P_{t}}\quad\lbrack{dB}\rbrack} = {{- 70} + {10{\log_{10}\left( {\Delta\quad L} \right)}} - {\frac{\Delta\quad Z}{100}.}}} & (41)\end{matrix}$

b. Laser Phase Locking Means.

Refer to FIG. 3. Local oscillator laser, 45, generates a localoscillator lightwave. The local oscillator lightwave propagates fromlocal oscillator laser, 45, to optical coupler or beamsplitter, 43. Theoptical coupler, 43, splits off the smaller portion of power of thelocal oscillator lightwave into optical pathway, 41. Optical path, 41,propagates the smaller portion of the local oscillator lightwave to thephase locking means optical receiver, 35. The larger portion of thepower of the local oscillator lightwave is split off by optical coupler43, and passed to optical path 13. Optical pathway, 13, propagates thelarger portion of the local oscillator lightwave to optical receiver,15. The phase locking means optical receiver, 35, receives andinterferes the transmitter laser lightwave from optical pathway, 39, andthe local oscillator lightwave from optical pathway 41. The receiver 35interferes the reference lightwaves from lasers 3 and 45 producing anelectrical output which is a radio frequency wave on electrical pathway,33. The electrical output, 33, provides an electronic beat frequencywhich directly indicates the difference in optical frequency and phasebetween lasers 1 and 45. Phase locking circuitry 31, employing aconventional phase lock loop mechanism, controls the difference infrequency between laser 1 and 45 and phase locks the two lasers to afixed frequency and phase relationship as indicated by the dashed linebetween circuitry 31 and local oscillator laser 45. The radian frequencydifference is Δω as discussed early in the text. The purpose of thelaser phase locking means is to insure that the local oscillatorlightwave traveling on optical path, 13, into optical receiver, 15, hasthe proper phase and frequency relationship to the composite lightwaveon optical pathway, 11. It is to be appreciated that the phase lockingmechanism also acts cooperatively with phase demodulator system 66 to bedescribed later herein. Conventionally, a common master clock oscillator311, FIG. 7 provides the timing base for both phase locking circuitry 31and an I & Q demodulator 300, FIG. 7.

Refer to FIG. 3. The composite lightwave on optical path 11, is an inputinto optical receiver 15. The local oscillator lightwave on opticalpath, 13, is also an input into optical receiver, 15. The localoscillator and composite lightwaves are interfered on photodetectorsproducing an electronic signal which electronically represents theheterodyned optical interference power between the two lightwaves. Theresulting composite radio frequency signal at output, 17, representselectronically the composite lightwave signal on optical path, 11. Thecomposite electronic receiver signal is passed from optical receiveroutput, 17, through amplifier, 19, via electronic path, 21, to thecorrelator system, 23. The local oscillator lightwave on optical path,13, is interfered with the composite lightwave on optical path 11. Theinterference power is photodetected in optical receiver, 15, byoptically interfering the composite back propagating lightwave on thelocal oscillator signal. As one of the components of this interferingaction, there is produced a difference beat signal which is a compositeradio frequency representation of the composite light wave on opticalpath, 11.

This interfering of the local oscillator output lightwave 13 and thecomposite back-propagating CW lightwave 11 provides the translation ofsignal 11 from the optical domain to a CW radio frequency (r.f.)composite difference beat signal 17. This reduces the frequency ofsignal 15 into an electronically processable signal frequency range. Itis to be appreciated that an important aspect of the present inventionthat the r.f. composite difference signal produce by this translationaction includes having counterpart components of the aforesaidcomponents of the composite back-propagating lightwave signal, with thephase states of these counterpart r.f. domain signals the same as thephase states of the corresponding components of the back-propagatinglightwave.

In accordance with the present invention, lasers 3 and 45 are to havesufficiently stringent high performance capability with respect toexactness of frequency to enable interference effects therebetween andheterodyne detection of acoustic perturbation signals incident to fiber9 to produce beat frequencies within the radio frequency (r.f.) range.Also in accordance with the present invention, lasers 3 and 45 havestringent performance criteria with respect to the phase stability, orcoherence, of their beams. They are to be substantially coherent over atleast a propagation path distance substantially equal to twice thelength, L, of sensing fiber 9. For example, a commercially availablenon-planar, ring laser (e.g. Lightwave Electronics Corp. Model 125)would be suitable for an intruder sensing perimeter intrusion monitoringfiber 9 having a length of 8.0 km (approximately 5 miles). The laserbeam of this commercially available laser, which is in the near infraredrange, has a frequency of 227 terahertz, or 1319 nanometer wavelength,and has a frequency stability accurately within one part in a billionover 1 millisecond period, or 5 Kilohertz in a 1 millisecond period.

It is to be appreciated that the provision of such frequency and phasestability of lasers 3 and 45 enables implementing the phase locking toproduce a sufficiently small non-zero radian phase locking circuitry 31.This in turn enables lasers 3 and 45, under regulation by phase lockingcircuitry 31, to provide a pair of beams which are phase locked and witha “non-zero Δω” sufficiently small to enable a heterodyne-mode opticalreceiver to provide the desired beat frequency outputs in the r.f.range. It is understood that laser 45, optical receiver 35, circuitry 31and beamsplitter 43 could be replaced with an apparatus applying thenon-zero ΔW to the beam from optical pathway 39 to give the same result.The returned interrogation optical composite wave is defined in thepreceding subsection 3(a) “Optical Transmitter and Time-delayMultiplexing Process” of this DESCRIPTION OF THE PREFERRED EMBODIMENT.

In the preceding section (1) “Description of Underlying Theories” ofthis DESCRIPTION OF THE PREFERRED EMBODIMENT there is a definition of“non-zero Δω” and a mathematical demonstration of its importance in theheterodyne mode of interfering. It makes it possible to use relativelysimple processes to avoid fading. By way of contrast, fading with the“zero Δω” homodyne mode of interfering would entail much more difficultand less effective fade avoidance processes.

c. Correlation Time-Delay Demultiplexing.

Refer to FIG. 3. The composite radio frequency signal on electricalpath, 21, is input into the correlator system, 23. The correlator systemdelays the master correlation code generator output, 51, an appropriateamount and correlates the delayed correlation code with the compositeradio frequency signal. This produces electrical outputs O₁, O₂ . . .O_(n) corresponding to signals S₁, S₂ . . . S_(n), in turn correspondingto spatial delays L₁, L₂ . . . L_(n). The spatial delays L₁, L₂ . . .L_(n) are arbitrary and programmable. The electrical output O₁corresponds to B(t,L₁) referred to in the preceding subsection 2(a).

The correlation process is well understood in the literature. The signalthat represents the backscattered optical wave in array, 9, that ispassed from the optical receiver 15, to the correlator system 23,contains all of the information for all sensors or channels S₁, S₂ . . .S_(n) at once on the electronic signal path 21 entering the correlator23. Because the backscattered composite signal is modulated with thecorrelation code by modulator 5, the backscattered light is timestructured with the time structure of the correlation code. Because thecorrelation code is selected to have special autocorrelation codeproperties, the time structure of the correlation codes allows anelectronic representation of the backscattered light at positions L₁, L₂. . . L_(n) to be obtained via the correlation process in the correlator23. In a preferred embodiment of the invention the master code generator53 is a shift register type pseudorandom number (PRN) code generator andeach correlator of the set 23 would be a correlation type demodulatorherein later described in greater depth. Code generator 53 mayalternatively be embodied as a binary sequence having transorthogonalautocorrelation properties (binary pseudonoise sequence) and eachcorrelator would then be a correlation-type demodulator for demodulatinga binary pseudonoise sequence, whose implementation would be understoodby those of skill in the art. The correlator uses the referencecorrelation code from correlation code generator, 53, which is passedvia electronic path 51 to the correlator, 23, as a “golden ruler”enabling sorting out by temporal and spatial domain demultiplexingelectronic representations of the backscattering optical signals atsensors or channels S₁, S₂ . . . S_(n). Various delayed versions of thecorrelation code are multiplied by the composite signal with all of thesensor or channel signals present simultaneously, from electronic path21 so that the electronic representations of the sensors or channels S₁,S₂ . . . S_(n) are output from the correlator, 23 on signals O₁, O₂ . .. O_(n) with respect to the index.

Correlator system 23 is an electronic spread spectrum signal de-spreaderand temporal and spatial domain de-multiplexer of the r.f. signalcounterpart to the optical composite signal. Its input is coupled to theamplified output 21 of the heterodyner and photodetector, and it isoperative in cooperation with said source of spectrum spreading signalsto perform a coherent signal correlation process upon the r.f.counterparts of the aforesaid “one” and the aforesaid “still another”components of the composite back-propagating CW lightwave. This causesthe de-spreading of the r.f. counterpart of the optical reflected spreadspectrum signal and causes the temporal and spatial demultiplexing ofthe r.f. counterpart of the “still another” component of the compositer.f. signal. This processing provides signals which temporally andspatially sort the said “still another” component into n virtual sensorsignal channels, or stated another way n of each of the ΔZ lengthmeasurement regions, measuring the induced optical path change at eachof the n ΔZ-length segments of the optical fiber span 9.

It will be appreciated that this sorting process is accomplished by theautocorrelation properties of the spectrum-spreading signal and by thetime of flight of the optical spectrum-spreading signal down to each nthreflection segment and back to the heterodyne optical receiver 15. Adelayed replica of the spectrum-spreading signal is correlated againstthe r.f. signal counterpart of the optical composite back-propagatingsignal, thereby segmenting the optical fiber into n independentsegments, or virtual sensors, via the time of flight of the opticalcomposite back-propagating signal and the autocorrelation function ofthe transmitted spectrum-spreading signal.

It is to be appreciated that system 2 is operating in the spreadspectrum transmission and reception mode. Namely, by providing opticalinterrogation light wave, E₁(t), with modulation by the correlationcode, c(t), the continuous wave carrier signal is temporally structuredinto a spread spectrum interrogation lightwave which continuouslyreiterates autocorrelatable code sequences. Then after correlationsystem provides an appropriate time of delay the correlator system 26correlates the backscattered light wave E_(b)(t) with the same output,c(t), of code generator 53, de-spreading the spread spectrum signal.

In accordance with well known communication electronics theory this hasthe effect of increasing signal output of the ROSE sensor system whilethe noise bandwidth remains the same. In temporally and spatiallysorting the r.f. counterpart of the aforesaid “still another” componentof the composite back-propagation lightwave, the aforesaid “another”component of undesired noises, such as reflections from couplers infiber span 9, are materially attenuated.

More particularity, in accordance with this well known theory, thesignal-to-noise ratio (SNR) is enhanced by considerable attenuation ofnoise mechanisms in frequency ranges outside of center frequency lobe ofthe autocorrelation function and outside the pair of first side lobes toone and the other side of the center frequency lobe.

An illustrative embodiment of electronic spread spectrum signalde-spreader and spatial de-multiplexer for cooperation with thepreviously described shift register type PRN code generator may comprisea series of n like-shift register code generators respectively receivingthe spectrum spreading signal through a corresponding series of n feedchannels which cause delays which incrementally increase by an amount oftime bearing a predetermined relationship to the fiber span length, andC_(L), the speed of light through the fiber. The composite r.f. signalis fed to a corresponding series of n multipliers connected to receiveas the other multiplier the codes generated by the respectivede-spreader and demultiplexer to thereby provide the de-spread andde-multiplexed signal.

d. Heterodyne Phase Demodulation.

Refer to FIG. 3. After the composite radio frequency signal onelectrical path 21 is correlation time-delay demultiplexed by thecorrelator system, 23, the plurality (which upwardly may include a verylarge number, for instance 5,000) of outputs O₁, O₂ . . . O_(n), on theplurality of electrical paths 61, 63 and 65 respectively are phasedemodulated by a plurality of individual phase demodulations in thephase demodulator system, 66. The outputs of the phase demodulatorsystem, 66, are the corresponding plurality of electrical paths 71, 73,and 75. The phase demodulator outputs 71, 73, and 75 correspond to thecorrelator outputs (O₁, O₂ . . . O_(n)) 61, 63 and 65 respectively, andto the corresponding plurality of corresponding signals S₁, S₂ . . .S_(n) respectively corresponding to spatial delays L₁, L₂ . . . L_(n)respectively. The outputs 71, 73, and 75 electronically indicate (withtens of kilohertz potential bandwidth) the phase states of opticalsignals S₁, S₂ . . . S_(n). In particular, output 71 is proportional tothe temporal phase Φ₁ of B(t, L₁) hereinbefore discussed in subsections1(b) “Correlation or Time-delay Multiplexing” and 3(c) “CorrelationTime-Delay Demultiplexing” of this DESCRIPTION OF THE PREFERREDEMBODIMENT. The phase demodulator outputs 73 and 75 indicate thetemporal phase states Φ₂ and Φ_(n) of B(t,L₂) and B (t,L_(n))respectively.

e. Fading Free Polarization Processing

Preferably system 2 further includes polarization signal characteristicprocessing functions (not shown), which are used together with thepreviously described feature that the heterodyning function provides inreducing fading, of the backscattering signal, E_(b)(t). Thesepolarization processing functions are disclosed in the commonly assignedU.S. Pat. No. 6,043,921 entitled “Fading-Free Optical Phase RateReceiver,” hereby incorporated herein in its entirely. The opticalheterodyning feature which provides benefit in reducing fading includes:(i) cooperation of phase locked lasers 3 and 45 in the formation of theoptical interrogation light wave, E₁(t), applied to optical fiber 9, orother linearly extended light propagation medium producing Rayleigheffects backscattering, and (ii) the manipulation of this by opticalreceiver 15 to provide the composite electronic receive signal asoptical receiver output 17. This takes advantage of the feature of morefavorable Heterodyne fading conditions in a way, in which polarizationand phase state signal fading is materially reduced in the detectedbackscattered light wave E_(b)(t). The electronic decoding module 700 ofU.S. Pat. No. 6,043,921 is substantially an equivalent to the correlatorsystem herein. However, the system disclosed in U.S. Pat. No. 6,043,921for implementing polarization fading reduction (if not substantiallyeliminating fading) is a generalized stand alone system for processingany optical phase signal having temporally varying polarization, phase,and phase frequency. It must be adapted for application to system 2 byappropriate integration into system 2 included the two followingalternative approaches.

One approach for such adaptation passes the fade-free optical phase rate(FFOPR) photoreceiver RF signal to the correlator 23, performs thecorrelation on the RF signal and completes the Phase Demodulation by Inphase and Quadrature phase (hereinafter I & Q) demodulating thecorrelated RF signal into outputs. This method creates low bandwidth I &Q components and therefore requires low bandwidth analog-to-digitalconverters (implying a requirement for a large number of analog RFcorrelation electronic components). This RF correlator approach requirestwo correlator circuits for every virtual sensor element, or spatialchannel, along fiber 9. One correlator is needed for the verticalpolarization RF signal path and one correlator is needed for thehorizontal polarization RF signal path.

Another approach applies the I & Q demodulator of FIG. 7 of the U.S.Pat. No. 6,043,921 prior to correlation. This approach thereforecorrelates a wideband set of four I & Q signals. One I, Q, set is forhorizontal polarization and the other I, Q, set is for the verticalpolarization. In this case the I & Q signals are the I & Q signals forthe whole virtual array rather than for one virtual sensor element ofthe array. Four correlators are required for each sensor element. Onecorrelator is applied to each of the four wide bandwidth I & Q signalsfor each virtual sensor element. This second approach requires verywideband analog-to-digital converters, but allows digital correlators tobe used instead of analog RF correlators. The RF correlator or firstapproach requires far more analog to digital converters and RFelectronics. The digital correlator approach enables the correlators tobe implemented by the digital approaches of massively integrated logiccircuits and/or programmed processors, requiring far more digital logic,but substantially reducing the r.f. electronics and numberanalog-to-digital converter units in the system.

f. Phase Differencing

Refer to FIG. 3. The plurality (which upwardly may include a very largenumber, for instance 5,000) of signals indicating the phase states Φ₁,Φ₂ . . . Φ_(n) on electrical paths 71, 73 and 75, respectively, areinput into the phase differencer, 99. The phase differencer forms acorresponding plurality of outputs 91, 93 and 95 which are arbitrarilyand programmably assigned as the subtractions of any two pairs of phasesignals Φ_(j) and Φ_(k) (where j and k are selected from 1, 2 . . . n).

Each of the programmably selectable pairs of differenced phase signalsform a signal ΔΦ_(kj) which is spatially bounded within the region ofthe fiber between lengths L_(j) and L_(k). The phase differencertherefore produces differential phase outputs corresponding to a set ofprogrammable length and position virtual sensors.

Stated another way, each programmable selection of pairs of phasesignals forms a virtual spatial differential sensor which senses thedifference between the phases of the Δω output of the photodectorsub-system (which is the subject of the next subsections) in receiver15. Each Δω is an r.f. difference beat signal representative of theaforesaid “still another” component of the composite back-propagating CWlightwave signal which passes from the launch end of fiber span 9 todirectional coupler 7. These signals from each pair therefore representsignals of virtual spatial differential sensors along fiber span 9. As aresult of the choice of pairs being selectively programmable thesevirtual sensor can be employed to implement adaptive apertures inprocessing signal incident the fiber span. This feature would be useful,for example, in enabling security system operators to classify objectscausing acoustic pressure wave signals incident up a fiber span 9 usedas a perimeter intrusion monitoring line.

g. Optical Detector Sub-System.

The optical receivers 15 and 35, FIGS. 3, 4 and 5, are comprised ofphotodetector sub-systems. Any of the many well known photodetectingtechniques and devices may be employed. Possible implementation of thephotodetection sub-systems will now be discussed.

Refer to FIG. 4. Like parts correspond to like numbers. Optical signalsenter the photodetector sub-system via optical paths 101 and 103 whichare extensions of the paths 11 and 13 in the case of receiver 15, and(not shown) of paths 39 and 44 in the case of subsystem 35. The opticalsignals are equally split by optical coupler or beamsplitter, 105. Theoptical signal on path 107 is composite signal comprised of half theoptical power of path 101 and half of the optical power arriving on pate103. The optical signal on path 107 is illuminated on optical detector111. The photo-current of optical detector 111 flows into electricalconductor 115. Likewise, the optical signal on path 109 is comprised ofhalf the optical power on path 101 and half of the optical power on path103. The optical signal on path 109 is illuminated on optical detector113. The photo-current of optical detector 113 flows out of electricalconductor 115. Therefore the photo-currents of optical detectors 111 and113 are subtracted at electrical conductor or node 115.

Photo-detectors 111 and 113 are precisely matched in responsivity. Thedifferential photocurrent on electrical conductor 115 is input intopre-amplifier 117, amplifier and is passed to electrical output 119. Thedifferential nature of the photo-detection rejects either of theself-optical interference power of the signals on paths 101 and 103 andreceives only the cross-interference power between the two opticalsignals on paths 101 and 103. This particular optical detectorarchitecture is called a balanced heterodyne optical detection scheme.The scheme is 3 dB more sensitive than all other heterodyne opticaldetection methods and offers the distinct advantage of rejecting localoscillator noise.

Refer to FIG. 5. FIG. 5 illustrates an alternative photo-detectionscheme to FIG. 4. Lightwaves enter the receiver at paths 101 and 103.The optical coupler or beamsplitter 105 combines the lightwaves on paths101 and 103 into a composite lightwave on path 107. The compositelightwave on path 107 illuminates optical detector 111. Thephoto-current of optical detector caused by the self-interference andcross interference of lightwaves originating from optical paths 101 and103 passes through conductor 115 a, is amplified by pre-amplifier 117and is passed to electrical output 119.

The optical detector sub-system of FIGS. 4 and 5 correspond to opticalreceivers 15 or 35 of FIG. 3. Paths 101 and 103 correspond to 11 and 13and output 119 corresponds outputs 17 in optical receiver 15. Paths 101and 103 correspond to 39 and 41 and output 119 corresponds to output 33in optical receiver 35. Either of the photo-detection schemes of FIG. 4or 5 can be used for the optical receivers 15 or 35. However, thephotodetection scheme of FIG. 4 is preferred.

h. Programmable Correlator System

Refer to FIG. 6. The composite radio frequency signal, or r.f. compositereference beat signal, which electronically represents the receivedtime-delay multiplexed optical signal, or composite back-propagation CWlightwave, E_(b)(t), is input into the correlator system, 23, atelectrical input 21. The composite radio frequency signal is n-way splitwith power splitter 203 into a plurality (which upwardly may include avery large number, for instance 5,000) of electronic pathways including211, 213 and 215. The master correlation code, c(t), is input into thecorrelator system, 23, at electrical input 54. The correlation code isdistributed to such a plurality of programmable delay circuits including221, 223 and 225. Each programmable delay circuit delays the mastercorrelation code by the delay required to decode/demultiplex eachtime-delay multiplexed channel. The plurality of programmable delaycircuits including 221, 223 and 225 output a plurality of delayedcorrelation codes including those on electrical pathways 231, 233, and225 respectively. The corresponding plurality of delayed correlationcodes including those on electrical pathways 231, 233 and 235 aremultiplied by a corresponding plurality of multipliers (or balancedmixers) including 241, 243 and 245, respectively, by the radio frequencysignal on the plurality of electronic pathways including 211, 213 and215 which are amplified by a corresponding plurality of amplifiersincluding 261, 263 and 265, respectively, to produce the correspondingplurality of outputs including O₁, O₂, and O_(n) (on lines 61, 63 and65) respectively. Each of the outputs therefore produces thecorresponding demultiplexed signal which is time-gated by thecorresponding time-delay of the correlation code. The correlator system23 of FIG. 6 is an example implementation of the correlation system, 23,of FIG. 3.

The output O₁ corresponds to signal B(t,L₁) which is hereinbeforediscussed in subsections 2(a) “ROSE Optical Phase Sensor InterrogationEnables Sensor System” and 3(c) “Correlation Time-Delay Demultiplexing”of this DESCRIPTION OF THE PREFERRED EMBODIMENT. The output O₁, O₂ . . .O_(n) on lines 61, 63 and 65, respectively, correspond to signals S₁, S₂. . . S_(n) which in turn are based upon the spatial delay associatedwith distance L₁, L₂ . . . L_(n) indicated in FIG. 3. These spatialdelays are based on the time of propagation for flyback travel alongthese distances, which are arbitrary and programmable. The time delaymultiplexing of the optical signals comprising the compositeback-propagating optical signal on path 11, FIG. 3, arise from aplurality (which upwardly may include a very large number, for instance5,000) of spatial locations causing a like plurality of time-delays. Thecorrelator system spatially separates the components of the r.f.composite difference beat signal into channels which each uniquelyrepresent an optical signal at a single spatial location.

The correlator system allows the spatial sampling of the optical signalsso that a virtual array can be formed along the fiber span 9 on FIG. 3.

i. Phase Demodulation System The embodiment of phase demodulator system,66, of FIG. 3, has two uses in system 2. It either: (i) receives theoutputs of the just described r.f. correlator subsection 23, or (ii) ispart of the integration of the polarization fading reduction system ofU.S. Pat. No. 6,043,921 (as discussed in the preceding subsection 2(e)“Fading-Free Polarization Processing” of this DESCRIPTION OF THEPREFERRED EMBODIMENT. Refer to FIG. 7. The phase demodulation system,66, is comprised of a plurality (which upwardly may include a very largenumber, for instance 5,000) of phase demodulators, 81, 83 and 85. Theinputs to the plurality of phase demodulators, 61, 63 and 65 (thecorrelator outputs O₁, O₂ . . . O_(n) discussed previously) are phasedemodulated with phase demodulators 81, 83 and 85 respectively. Theoutputs of these demodulators are passed on electrical pathways 71, 73and 75 respectively.

Refer to FIG. 8. An example block diagram of any one of the justdiscussed phase demodulators 81, 83 and 85 is shown as part 300. Theinput electrical path 301 corresponds to any one of electrical path 61,63, 65, etc. of the plurality of phase demodulators. The outputelectrical path 319 corresponds to any one of electrical path 71, 73,75, etc. of the plurality of phase demodulators. A correlation systemoutput such as O₁, O₂ or O_(n) is passed via electrical path 301 into abandpass filter 303. The bandpass filter 303 passes only a band ofradian frequencies in the vicinity of Δω so that only B(t,L_(m)) passesthrough the filter (where m is an integer corresponding to theparticular channel). The band passed signal passes from 303 viaelectrical path 305 to amplitude control 307. Amplitude control 307 iseither an analog automatic gain control circuit, an electronic clippercircuit, or a combination thereof. The amplitude control 307 removesamplitude variations due to polarization fading or other types of signalfading. Because the signal, B(t,L_(m)) is a result of a heterodyneinterference, the phase remains the same after clipping. It is to beappreciated that other phase demodulation schemes for fiber opticsignals use a phase carrier technique which does not allow the clippingoperation. Clipping is a preferred amplitude control mechanism. Theamplitude control 307 passes an amplitude stabilized signal viaelectrical path 309 to I & Q demodulator 311. The I & Q demodulatorremoves the carrier, that is it shifts the center radian frequency ofthe amplitude stabilized B(t,L_(m)) from Δω down to zero. The I & Qdemodulator outputs a voltage proportional to cos(Φ_(m)) on electricalpath 313 and a voltage proportional to sin(Φ_(m)) on electrical path315. The cos(Φ_(m)) and sin(Φ_(m)) proportional voltages on electricalpaths 313 and 315 respectively are converted in an output signalproportional to (m on electrical path 319 by the phase detector 317.

Reviewing the previous discussion, the plurality of phase demodulators81, 83 and 85 of FIG. 7 each function like the block diagram of 300 onFIG. 8. The plurality (which upwardly may include a very large number,for instance 5000) of phase demodulators 300 convert to a like pluralityof signals B(t,L₁), B(t,L₂) . . . B(t,L_(n)) into a like plurality ofsignals proportional to Φ₁, Φ₂ . . . Φ_(n) which correspond to opticalsignals S₁, S₂ . . . S_(n).

j. I & Q Demodulator.

An example implementation of the I & Q demodulator 311 of FIG. 8 willnow be presented. Refer to FIG. 9. An amplitude stabilized B(t,L_(m))signal (originating from the amplitude control 307 of FIG. 8) is passedon electrical path 309 to a power splitter 403. Half of the signal powerexiting from power splitter 403 is passed to analog mixer, balancedmixer, Gilbert cell or analog multiplier 413 via electrical path 411.The other half of signal power exiting form power splitter 403 is passedto analog mixer, balanced mixer, Gilbert cell or analog multiplier 423via electrical path 421.

The reference oscillator 451 generates an electronic wave proportionalto cos(Δωt). As noted earlier herein, this reference oscillator is alsothe oscillator employed in the conventional phase lock mechanismestablishing the fixed phase relationship between the frequencies ofprimary laser 3 and local oscillator laser 45 whose differences infrequency, ΔW, are of a very low order. In accordance with knownprinciples of heterodyning lightwaves having fixed phase relationships,heterodyning these signals can produce a difference beat signal smallenough to be in the r.f. signal range, but with the frequency differencesufficiently high to provide the heterodyning with a band pass allowingtransforming a given binary code rate into corresponding code componentsof the beat signal, such as the code rate of the PRN code sequenceproduced by PRN code generator 53. This reference oscillator wave ispassed from the reference oscillator 451 via the electrical path 453 toamplifier 455. The wave is amplified by amplifier 455 and passed tohybrid coupler 459 via electrical path 447. The hybrid coupler splitsthe amplified reference oscillator electronic wave into two componentsone proportional to cos(Δωt) on electrical path 417 (providing the “I”,or In-phase reference); and one proportional to sin(Δωt) on electricalpath 427 (providing the “Q”, or Quadrature-phase reference).

The In-phase reference on electrical path 417 is multiplied (orfrequency mixed) with the signal on electrical path 411 by multiplier413 to produce the output on electrical path 415. The signal onelectrical path 415 is amplified by amplifier 431 and passed toelectronic lowpass filter 435 via electrical path 433. The lowpassfilter 435 removes high frequency components of the multiplication orfrequency mixing process and results in an output at electrical path 313which is proportional to cos (Φ_(m)).

The Quadrature-phase reference on electrical path 427 is multiplied (orfrequency mixed) with the signal on electrical path 421 by multiplier423 to produce the output on electrical path 425. The signal onelectrical path 425 is amplified by amplifier 441 and passed toelectronic lowpass filter 445 via electrical path 443. The lowpassfilter 445 removes high frequency components of the multiplication orfrequency mixing process and results in an output at electrical path 315which is proportional to sin(Φ_(m)).

k. Phase Detector

Example implementations of the phase detection 317 of FIG. 8 will now bepresented. Refer to FIG. 10. An example digital phase detectorimplementation, 317, is shown in the block diagram. The signalproportional to cos (Φ_(m)) on electrical path 313 is converted to adigital code or number by analog-to-digital converter (hereafter, A/D)513. The digital number proportional to cos (Φ_(m)) is input into thedigital signal processor 501 via electrical path 515. The signalproportional to sin (Φ_(m)) on electrical path 315 is converted to adigital code or number by A/D 523. The digital number proportional tosin(Φ_(m)) is input into the digital signal processor, 501, viaelectrical path 525. The digital signal processor converts the numbersproportional to sin (Φ_(m)) and cos (Φ_(m)) into a number proportionalto Φ_(m) as follows.

Suppose the constant of proportionality for the sin(Φ_(m)) andcos(Φ_(m)) is V_(m). Then the digital signal processor can optimallyselect estimates of Φ_(m) and V_(m) to minimize the calculated errorfunction:ε({circumflex over (Φ)}_(m) ,{circumflex over (V)} _(m))=((V _(m)cos(Φ_(m))−{circumflex over (V)} _(m) cos({circumflex over(Φ)}_(m)))²+(V _(m) sin(Φ_(m))−{circumflex over (V)} _(m) sin(Φ_(m)))²)  (42)

The digital signal processor can also calculate Φ_(m) directly by takingthe inverse tangent function or the inverse cotangent function:$\begin{matrix}{\Phi_{m} = {{a\quad{\tan\left( \frac{V_{m}{\sin\left( \Phi_{m} \right)}}{V_{m}{\cos\left( \Phi_{m} \right)}} \right)}} = {a\quad{\cot\left( \frac{V_{m}{\cos\left( \Phi_{m} \right)}}{V_{m}{\sin\left( \Phi_{m} \right)}} \right)}}}} & (43)\end{matrix}$

If desired, the digital signal processor can also implement thedifferentiate and cross multiply (hereafter DCM) algorithm. The DCMmethod is as follows. The digital representation of the signalsproportional to sin (Φ_(m)) and cos (Φ_(m)) are temporallydifferentiated and cross multiplied by the non-differentiated signals.The result U_(m)(t) is integrated to produce the desired output, Φ_(m).Mathematically, this algorithm is: $\begin{matrix}\begin{matrix}{{U_{m}(t)} = {{V_{m}{\sin\left( \Phi_{m} \right)}\frac{\partial}{\partial t}\left( {V_{m}{\cos\left( \Phi_{m} \right)}} \right)} - {V_{m}{\cos\left( \Phi_{m} \right)}\frac{\partial}{\partial t}\left( {V_{m}{\sin\left( \Phi_{m} \right)}} \right)}}} \\{{U_{m}(t)} = {{V_{m}^{2}\left( {\left( {\cos\left( \Phi_{m} \right)} \right)^{2} + \left( {\sin\left( \Phi_{m} \right)} \right)^{2}} \right)}\frac{\partial\Phi_{m}}{\partial t}}} \\{{U_{m}(t)} = {V_{m}^{2}\frac{\partial\Phi_{m}}{\partial t}}} \\{\Phi_{m} = {\frac{1}{V_{m}^{2}}{\int{{U_{m}(t)}{{\partial t}.}}}}}\end{matrix} & (44)\end{matrix}$The digital signal processor 501 converts the signals arriving onelectrical paths 515 and 525 into a digital output proportional to Φ_(m)on electronic path 503. Optionally, the digital output is passed onelectronic path 505 to some other data sink such as a computer memory.The digital signal proportional to Φ_(m) on electronic path 503 isconverted back to an analog signal on electrical path 319 bydigital-to-analog converter 507. By way of a summarization, the exampledigital phase detector 317 accepts inputs 313 and 315 which originatefrom the I & Q demodulator, 311, of FIG. 8, and the digital phasedetector 317 outputs the phase signal Φ_(m) on electrical path 319.Optionally, any of other well-known implementations of digital phasedetectors may be employed.

Refer to FIG. 11. An example analog phase detector implementation, 317′is shown in the block diagram. The example analog phase detector 317′shown in FIG. 11 implements an analog version of the DCM algorithmdiscussed in the previous text. The signal proportional to cos (Φ_(m))on electrical path 313 is input into analog temporal differentiator 613and analog multiplier 617. The signal proportional to sin(Φ_(m)) onelectrical path 315 is input into analog temporal differentiator 623 andanalog multiplier 627. The differentiated cosine term on signal path 625is multiplied by the sine term on electrical path 315 by analogmultiplier 627 producing the signal on electrical path 629. Thedifferentiated sine term on electrical path 615 is multiplied by thecosine term on electrical path 313 by analog multiplier 617 producingthe signal on electrical path 619. The signals on electrical paths 619and 629 are applied as inputs to differential summer 631. The output ofdifferential summer on electrical path 633, which is the result of thedifferentiated sine and cosine product being subtracted from thedifferentiated cosine and sine product, corresponds to U_(m)(t) of theDCM discussion. The signal on electrical path 633 is integrated byanalog integrator 635 to produce the analog phase detector outputproportional to Φ_(m) on electrical path and output 319. By way ofsummarization, the example analog phase detector 317 accepts inputs 313and 315 which originate from the I & Q demodulator 311 of FIG. 8, thenthe analog phase detector outputs the phase signal Φ_(m) on electricalpath 319. Optionally, any of other well-known implementations of analogphase detectors may be employed.

1. Programmable Phase Difference

The example programmable phase differencer implementation shown as part99 of FIG. 12 corresponds to part 99 shown as a block in FIG. 3. Referto FIG. 12. The plurality (which upwardly may include a very largenumber, for instance 5,000) of demodulated signals proportional tooptical signal phases Φ₁, Φ₂ . . . Φ_(n) are input into the programmablephase signal switching and routing network 701 via electrical paths 71,73 and 75, respectively. Network 701 programmably selects on a basis oftimed relation to code generator 53 and routes on a basis ofconventional “hold-in memory” and “transfer-from-memory”, a plurality(which upwardly may include a very large number, for instance 5,000) ofpairs of phase signals onto a plurality (which upwardly may include avery large number, for instance 5,000) of pairs of electronic paths 711and 713, 731 and 733 and 751 and 753. The plurality of routed pairs ofphase signals are applied to the corresponding of subtracters 715, 735and 755 as shown on FIG. 12. The plurality of phase pairs on electronicpairs of paths 711 and 713, 731 and 733, and 751 and 753 are subtractedby subtracters 715, 735 and 753, respectively, and the differentialsignal are outputted on a corresponding plurality of electrical paths91, 93 and 95 respectively. The following description focuses on thedifferencing channel output on electrical path 91, it being understoodthat the modes of operation of other differencing channels in network701 are the same. Programmable phase switching and routing network 701selects one of the phase signals on one of the plurality of electricalpaths 71, 73 or 75 and routes the signal to electrical path 711. Thesignal on electrical path 711 is selected to be proportional to Φ_(j)(where j is of the set 1, 2 . . . n). Network 701 also selects anotherof the phase signals on one of the other of the plurality of electronicpaths 71, 73 or 75 and routes the signal to electrical path 713. Thesignal of electrical path 713 is selected to be proportional to Φ_(k)(where k is of set 1, 2 . . . n). The signal on electrical path 711 issubtracted from the signal on electrical path 713 by subtracter 715. Theoutput of subtracter 715 is passed on via electrical path 91 and isproportional to ΔΦ_(kj) hereinabove discussed in subsection 3(f) “PhaseDifferencing” of this DESCRIPTION OF THE PREFERRED EMBODIMENT. Employingthis mode, network 701 programmably makes selection from optical signalphases Φ₁, Φ₂ . . . Φ_(n) to provide other differential phase outputs onelectrical paths 91, 93 and 95. This may include a very large number ofdifferential phase signals, for instance 5000. As an alternative to thejust described type of circuitry employing subtractors 715, 735 and 755any of other well-known forms of producing a differential signal my beemployed.

m. An Alternative Viewpoint of the Partitioning of System 2.

As an alternative to the viewpoint inferable from the preceding sequencediscussing FIG. 3, system 2 may be considered as partitioned into: (i)an optical network for illuminating an optical fiber sensing span, orother light propagation medium sensing span, and retrieving backpropagating portions of the illumination; and (ii) a photoelectronicnetwork for establishing virtual sensors at predetermined locationsalong the span and picking up external physical signals incident to, orimpinging upon, the sensors.

In general, the optical network for the illumination of, and for theretrieval of back-propagation from, fiber span 9 comprises transmitterlaser 3, directional optical coupler 7, and optical fiber, or otherlight propagation medium 9.

The photoelectronic network for establishing virtual sensors and pickingup signals therefrom generally comprises two subdivisions. Onesubdivision provides a cyclically reiterative autocorrelatable form ofmodulation of the lightwave illuminating fiber span 9. This modulationis in the form reiterated sequences having autocorrelatable properties.The other subdivision takes the retrieved back propagation and performsa heterodyning therewith to obtain an r.f. beat signal. It then picks upthe signal from the virtual sensors by autocorrelation and furtherprocesses it into more useful forms.

In general, the subdivision providing the cyclical reiterativemodulation of sequences illuminating fiber span 9 comprises mastercorrelation code generator 53 (via one of its electrical pathwayoutputs) and optical modulator 5.

In general, the subdivision for performing heterodyning with and pickingup of virtual sensor signals from the retrieved back propagation fromfiber span 9 includes local oscillator laser 45, and the network whichphase locks transmitter laser 3 and local oscillator 45, and a sequenceof elements which perform processing upon the retrieved backpropagation. The phase locking network comprises beamsplitter 4, phaselocking means optical receiver 35, phase locking circuitry 30, andoptical coupler 43. First in the sequence of processing elements is anoptical receiver 15 which photodetects interference power “derived” byheterodyning the back propagated illumination portion retrieved fromfiber span 9 with the output of a local oscillator 45. Lasers 3 and 45are operated with a frequency difference to produce an r.f. beat signal,ΔW. Then correlation system 23 receives as one of its inputs anotherelectrical pathway output from master correlation code generator 53, andprovides a series of channels which in turn respectively providepredetermined time delays in relation to the timing base of cyclicreiterative code generator 53, to perform a series of autocorrelationsof the respectively delayed inputs from code generator 53 with thesignal ΔW. This picks up r.f. signals respectively representative of theaffects in the lightwave domain of the external physical signalsincident upon the respective virtual sensor. Phase demodulator system 66provides a linear phase signal derived from such r.f. signalsrepresentative of optical signals at the respective virtual sensors.Programmable phase differencer 99 processes pairs of these linear phasesignals occurring across segments of fiber span 9 between programmablyselected pairs of the virtual sensors.

Following is another overview description which more particularly callsattention to an aspect of the invention that the system elements whichperform the autocorrelation enable providing an output in the form of anr.f. counterpart of a lightwave time-domain reflectometry output ofsignals incident to the virtual sensors as lightwave time domainreflectometry outputs A CW lightwave modulated by a continuouslyreiterated binary pseudorandom code sequence is launched into an end ofa span of ordinary optical fiber cable. Portions of the launchedlightwave back propagate to the launch end from a continuum of locationsalong the span because of innate fiber properties including Rayleighscattering. This is picked off the launch end and heterodyned to producean r.f. beat signal. The r.f. beat signal is processed by a plurality(which can be thousands) of correlator type binary pseudonoise codesequence demodulators respectively operated in different delay timerelationships to the timing base of the reiterated modulation sequences.The outputs of the demodulators provide r.f. time-domain reflectometryoutputs representative of signals (e.g., acoustic pressure waves)incident to virtual sensors along the fiber at positions correspondingto the various time delay relationships.

Following is still another overview description which more particularlycalls attention to an aspect of the invention that the system elementsperforming the autocorrelation enable detection of unique spectralcomponents representing a phase variations of external signals incidentto the virtual sensors. A CW lightwave modulated by a continuouslyreiterated pseudorandom code sequence is launched into an end of a spanof ordinary optical fiber cable. Portions of the launched lightwave backpropagate to the launch end from a continuum of locations along the spanbecause of innate fiber properties including Rayleigh scattering. Thisis picked off the launch end and heterodyned producing an r.f. beatsignal. The r.f. beat signal is processed by a plurality (which can bethousands) of correlator type pseudonoise code sequence demodulation andphase demodulator units, operated in different time delay relationshipsto the timing base of the reiterated modulation sequences. These unitsprovide outputs representative of phase variations in respective uniquespectral components in the r.f. beat signal caused by acoustic, or otherforms of signals, incident to virtual sensors at fiber positionscorresponding to the various time delay relationships.

Following is yet another overview description which more particularlycalls attention to an aspect of the invention that a pair of thedifferent delay time relationships of the autocorrelation systemelements are effective to establish a virtual increment of the opticalfiber span, and that a substracter circuit of phase differencer 99enables representing the differential phase signal across the virtualincrement. A CW lightwave modulated by a continuously reiteratedpseudorandom (PN) code sequence is launched into an end of a span ofordinary optical fiber cable. Portions of the launched lightwave backpropagate to the launch end from a continuum of locations along the spanbecause of innate fiber properties including Rayleigh scattering. Thisis picked off the launch end and heterodyned producing an r.f. beatsignal. The r.f. beat signal is processed by a plurality (which can bethousands) of correlator pseudonoise code sequence demodulation andphase demodulator units operated in different delay time relationshipsto the timing base of the reiterated modulation sequences. Pairs ofoutputs of the units are connected to respective substracter circuits,each providing a signal representative of signal differential ofincident acoustic signals, or other forms of signals, across virtualincrements of the span established by a pair of said delay timerelationships.

n. Air-Backed Mandrel Modified Form of Invention

FIG. 13 illustrates a so-called fiber-on-an-air-backed mandrel assembly801, useful in applications in which a fiber optic span 9′ is to beimmersed in a liquid medium. Assembly 801 comprises a hollow cylindricalmandrel 803 having formed therein a sealed central chamber 805containing air or other gaseous medium 807, which is compressiblerelative to the liquid medium. A segment of span 9′ of a ROSE system 2,FIG. 3, is helically wound the cylindrical exterior surface of mandrel803, and suitably fixedly bonded to the surface. The cylindrical wall809 of mandrel 803 is of a material so-chosen and of a thickness sochosen to form a containic membrane with a hoop stiffness that enablesacoustic pressure wave signals incident upon assembly 801 to betransformed into mandrel radial dimensional variations. As a result ofmandrel 801's geometry these radial variations result in magnifiedlongitudinal strain variations in fiber 9′. It is to be appreciated thatthe physical structure of assembly 801 inherently provides a spatialsuccession of two locations along the fiber span, which a phase signalswitch and routing network 701 could select and route to become thevirtual bounding positions of a differential phase signal virtualsensor. This is to say, positioning a mandrel wound span 9′ as a segmentof a system total span 9 of ROSE system 2 can facilitate providing asequential pair of virtual sensor locations along a span 9, and theprovision of a corresponding pair of delay circuits in correlatorcircuit 23 would cause assembly 801 to operate as a differential signalsensor.

(4) Advantages and New Features

The invention enables the interrogation or time-delay correlationalmultiplexing and demultiplexing of optical phase signals.

The invention enables the interrogation of ROSE (Rayleigh OpticalScattering and Encoding) fiber optic sensors. The invention enables thespatial sorting and separation of the temporal optical phases ofbackscattered optical signals arising from a plurality (which upwardlymay include a very large number, for instance 5,000) of virtual opticalsensors along fibers or other optical mediums. The invention enables thespatial decoding of backscattered optical signals with a bandwidth oftens of kilohertz. The invention enables the sensor locations along thefiber to be programmable. The invention allows the electronic separationor segmentation of the array of fiber sensors into programmable boundedlengths and positions. Because the correlation signal, c(t), can bedesigned to be a continuous wave, the invention increases the averageoptical power considerably over conventional pulsed optical phase sensorinterrogation methods. Because the correlation signal c(t) can be chosento have spectrum spreading properties for which dispreading electroniccircuitry is readily available, undesired optical fiber system noises,such as reflection discontinuity noises due to cable couplings, can bematerially attenuated.

In hypothetically assessing the potential achievable by the presentinvention with regard to employment of a common grade of optical fibercable buried beneath the ground surface as a perimeter intrusionmonitoring fiber span, the following assumptions have been made: (i)signal to noise ratio (S/N) degradation of Rayleigh effect lightpropagation in such an optical fiber cable are assumed to be 0.5 db/km;(ii) it is assumed there is a requirement for bandwidth of ten timesthat of the geo-acoustic intruder signal needs to be detected; (iii) anddigital circuitry functions are performed employing conventional “highend” clock rates. Using these assumptions, and employing conventionalsingle-mode or multimode fiber buried 6-12 inches underground, and usingconventional engineering methodology for noise effect prediction, it canbe shown that ROSE system 2 has the potential of sensing intruder causedgeo-acoustic, (i.e., seismic) signals along a length of fiber span lineas long as 8 km or 5 miles. (This assessment is based upon S/Ndegradations for flyback travel of signals from the interrogation launchend of fiber span 9 to its remote end and back.) The hypotheticalsegment resolution capability with such a 8 km, or 5 mile line, would be1 meter.

The invention provides a new capability of heterodyne optical phasedetection without resorting to dithered phase carrier methods. The phasedemodulation method introduces heterodyne I & Q demodulation to producecosine and sine phase components, clipped signal amplitude stabilizationtechniques and digital signal processing based phase detection. Thespatially differential phase detection method provided by the inventionenables the rejection of unwanted lead-in fiber phase signals.

The details, materials step of operation and arrangement of parts hereinhave been described and illustrated in order to explain the nature ofthe invention. Many modifications in these are possible by those skilledin the art within the teachings herein of the invention. For example,while in system 2 the transformation from optical to r.f. signal takesplace prior to processing by programmable correlation 23, it is withinthe skill of the art to design optical receiver 15 and correlator system23 to have the transformation take place otherwise. Also, as analternative to the previously described mechanism for phase lockinglaser 3 and 45, the laser optical wave on an optical path 39 can bepassed through an acoustic-optic modulator, sometimes called a BraggCell. The diffracted optical wave exiting the acousto-optic modulatorwill be Doppler shifted by an impinging-driving RF wave, that istranslated into a sound wave in the acousto-optic modulator, and theso-called Bragg shifted-diffracted optical wave will exit theacousti-optical modudulator with an optical frequency equivalent to thephase locked laser 45. The acousto-optically generated lightwave, at anequivalent frequency of the phase locked laser 45, is sent along opticalpathway 13 and becomes the local oscillator input to heterodynephotoreceiver 15. An acousto-optically frequency shifted version of thelight in optical path 39 can therefore replace the phase locked light ofcoherent optical source 45. Accordingly it is to be understood thatchanges may be made by those skilled in the art within the principle andscope of the inventions expressed in the appended claims.

1. A time-domain reflectometer for sensing at a desired set of n spaced sensing positions along an optical fiber span, said sensing positions being for sensing a type of external physical signal having the property of inducing light path changes within the optical fiber span at regions there along where the signal is coupled to the span, comprising: an optical fiber span having a first end which concurrently serves as both the interrogation signal input end and the back propagating signal output end for purposes of reflectometry, and having a second remote end; a first light source for producing a coherent carrier lightwave signal of a first predetermined wavelength; a spectrum spreading signal modulator for temporally structuring said carrier lightwave signal into a spread spectrum modulated interrogation lightwave signal which continuously reiterates sequences of an autocorrelatable spectrum spreading signal, the reiterated sequences being executed in a fixed relationship to a predetermined timing base; a light wave heterodyner having first and second inputs for receiving a primary signal and a local oscillator signal, respectively, and operative to produce the beat frequencies of their respective frequencies; a lightwave directional coupler having a first port which receives said spread spectrum modulated interrogation lightwave signal, a second port coupled to said first end of said optical fiber span, and a third port coupled to said primary signal input of the hetrodyner; said directional coupler coupling said spread spectrum modulated interrogation lightwave signal to said second port where it is launched in a forwardly propagating direction along said optical fiber span causing the return to said second port of a composite back-propagating lightwave signal which is a summation of the lightwave back-propagations from a continuum of locations along the length of the span, said composite back-propagating lightwave signal comprising a summation of multiple components including: a first signal component comprising the summation of portions of the said spread spectrum modulated interrogation lightwave signal which the innate properties of the optical fiber cause to back propagate at a continuum of locations along the span; and a second signal component comprising the modulation of said first signal component caused by longitudinal components of optical path changes induced into said span at a continuum of locations along said span by external physical signals, said second signal component further including a corresponding set of n subcomponents comprising the modulation of said first signal component by optical path changes caused by said external signals at the respective sensing positions; said directional coupler coupling said composite back-propagating lightwave signal to said third port where it is applied to said first input of the heterodyner; a second light source coupled to said second input of the lightwave heterodyner, said second light source producing a coherent local oscillator lightwave signal in phase locked relation to said carrier lightwave signal and of a second predetermined wavelength which differs from the first predetermined wavelength by an amount of difference small enough to produce at the output of the heterodyner a radio frequency (r.f.) composite difference beat signal, but by an amount large enough to cause said r.f. composite difference beat signal to have sufficient bandwidth to cause it to include r.f. counterparts of signal components and subcomponents of said composite back-propagating lightwave signal; said r.f. difference beat signal being coupled to an n-way splitter providing a corresponding set of n output channels, each transmitting said r.f. composite difference beat signal; a corresponding set of n de-spreaders and de-multiplexers having their respective inputs connected to the corresponding output channels of said n-way splitter through a corresponding set of time delay circuits which respectively provide a corresponding set of predetermined time delays in relation to said predetermined timing base of the spectrum spreading signal modulator, to establish said n desired sensing positions along said optical fiber span; and said set of r.f. de-spreaders and de-multiplexers concurrently serving multiple functions including: a first function of performing a coherent signal correlation process upon said r.f. composite difference beat signal to de-spread the r.f. counterparts of the interrogation lightwave signal; and a second function of conjunctively temporally and spatially demultiplexing said r.f. composite difference beat signal to provide at their respective outputs r.f. counterparts of the subcomponents of said second signal component of said composite back-propagating lightwave signal caused by changes in the optical path within said optical fiber span induced by external physical signals respectively coupled to the optical fiber span at corresponding sensing positions.
 2. The time-domain reflectometer of claim 1, wherein: said the multiple signal components of the composite back-propagating lightwave signal include an undesired third signal component due to back-propagations of the spread spectrum modulated lightwave interrogation lightwave other than caused by the innate properties of the optical fiber; and said first function of each r.f. de-spreader and de-multiplexer further serves the function of attenuating the r.f. counterpart of said undesired third signal component of said composite back-propagating lightwave signal.
 3. The reflectometer of claim 1, wherein: said innate properties of the said optical fiber material which cause back propagation include the generation of Rayleigh optical scattering effects at a continuum of locations along said optical fiber span in response to said forwardly propagating spread spectrum modulated lightwave interrogation signal.
 4. The reflectometer of claim 1, wherein said type of external physical signal which induces light path changes in said coupled to the optical fiber span is an acoustic pressure wave signal.
 5. The reflectometer of claim 4, wherein: said optical fiber span is an acoustic security alarm system perimeter monitoring line buried at a predetermined depth beneath the surface of the ground; said external physical signal being caused by the vibration of the ground surface by movement of an object thereon; and said set of n sensing positions along the line form a virtual array of n geophones which respectively produce signals representative of the vibrations of the surface at corresponding sensing positions.
 6. The reflectometer of claim 5, wherein: said optical fiber span is of a length L; and said first light source is a laser having the performance capability to generate a lightwave signal with sufficient phase stability to substantially retain coherency in propagation along said optical fiber span for a distance at least equal to 2 L.
 7. The reflectometer of claim 6, wherein: said the length L of said optical fiber span is at least 5.0 kilometers
 8. The reflectometer of claim 7, wherein said first light source is a planar, ring-type, laser.
 9. The reflectometer of claim 4, wherein said optical fiber span comprises a single-mode fiber optic cable.
 10. The reflectometer of claim 4 further comprising: a corresponding set of n phase demodulators for transforming each respective r.f. counterpart of each signal subcomponent of the set of n subcomponents of the second signal component of the composite back-propagating lightwave signal into a substantially linear signal representative of the radian phase of the corresponding lightwave subcomponent signal.
 11. The reflectometer of claim 10, wherein: said optical fiber span is buried beneath the surface of the ground as an acoustic security alarm system perimeter monitoring line; said set of n sensing positions along the span serving as perimeter line monitoring stations; and said outputs of said corresponding set of n phase demodulators provide substantially linear monitoring signals representative of ground vibrations caused by motion of objects on the surface of the ground near said optical fiber line.
 12. The reflectometer of claim 11, wherein the range of depths of burial of the optical fiber span beneath the surface of the ground is of the order of six inches to one foot.
 13. The reflectometer of claim 4, wherein said lightwave heterodyner is of the photodetector type.
 14. The reflectometer of claim 13, wherein: said lightwave heterodyner of the photodetector type is a balanced optical detector circuit including a matched pair of photodetectors with the composite back-propagating lightwave signal applied to each photodetector of the pair; and said balanced optical detection circuit produces said r.f. composite difference beat signal as a differential current from the matched pair of photodetectors.
 15. The reflectometer of claim 13, wherein each sequence of said continuously reiterated sequences of an autocorrelatable spectrum spreading signal is a binary pseudonoise code sequence of a type wherein shifts between binary states of the signal alternatingly shift the radian phase of the carrier between substantially 0° and 180°.
 16. The reflectometer of claim 15, wherein the pseudonoise code sequences are pseudorandom number (PRN) code sequences generated by a shift-register type PRN code generator.
 17. The reflectometer of claim 10 further comprising: a fixed frequency reference oscillator which produces a reference phase signal; each phase demodulator including an I & Q quadrature demodulator having a first input for receiving said reference phase signal and a second input for receiving an r.f. counterpart of the corresponding subcomponent of said second signal component of said composite back-propagating lightwave signal, said I & Q demodulator being operative to derive from said reference phase signal an interim in phase signal and an interim quadrature phase signal and to split the signal received at its second input, and mix one part thereof with the interim in phase signal, and mix another part thereof with the interim quadrature phase signal, to provide a pair of output signals; and each phase demodulator further including a phase detector having a pair of inputs for receiving respectively one and the other of said outputs of the I & Q demodulator and which is operative to provide at the output of the phase demodulator said signal representative of the radian phase of the corresponding lightwave subcomponent signal.
 18. The reflectometer of claim 1, wherein: a time period TP is required for forward propagation of said autocorrelatable spectrum spreading signal from the output of the source of the spectrum spreading signal to where said first light source is modulated, and then for the forward propagation of the derivative spread spectrum modulated interrogation lightwave signal to the second remote end of the fiber optical span, plus the time period required for the back propagation of a subcomponent of said second component of the composite back-propagating lightwave signal produced at the remote end of the span to the input of the heterodyner, and then for the propagation of the derivative counterpart subcomponent of the r.f. composite difference beat signal from the output of the heterdyner to the input of a corresponding de-spreader and de-multiplexer of said set of n de-spreader and de-multiplexers; and the temporal length of a single autocorrelatable spectrum spreading signal sequence of the continuously reiterated code sequences is one of one and the other of less than the time period TP, and greater than the time period TP.
 19. The reflectometer of claim 1, wherein said type of external physical signal which induces light path changes in said optical fiber span is a selected one of a group consisting of: (i) a seismic signal wherein with the media which couples the signal to said optical fiber span includes at least in part the ground in which the fiber optic span is buried; (ii) an underwater sound signal wherein the media which couples the signal to said optical fiber span includes at least in part a body of water in which the fiber optic span is immersed; (iii) an electromagnetic force field coupled to the optical fiber span; (iv) a signal comprising temperature variations coupled to the optical fiber span; and (v) at least one microphonic signal which is coupled to said optical fiber span at an at least one of said set of n sensing positions along the optical fiber span.
 20. The reflectometer of claim 4, wherein said optical fiber span comprises a multi-mode fiber optic cable.
 21. The reflectometer of claim 4, wherein said optical fiber span comprises a fiber optic cable of the polarization preserving type.
 22. The reflectometer of claim 4, wherein: said optical fiber span has a coating made of a thermoplastic material having the combined characteristics of a low Young's modulus and a Poisson's ratio below that of natural rubber; said coating serving to enhance the longitudinal. component of strain variation derived from an acoustic wave signal whose wave front is incident to the span from a direction at least in part having a lateral component parallel to the direction along which the wave front propagates.
 23. The reflectometer of claim 4, wherein at least one adjacent pair of the set of n sensing positions is intended for use immersed in a liquid medium wherein the media which couples said acoustic pressure wave signal at least in part includes said body of water, said reflectometer further comprising: there being provided in association with said at least one pair of said n sensing positions at least one mandrel having a cylindrical wall forming a central chamber therein; at least a portion of the optical fiber span between said at least one pair being helically wound around and fixedly bonded to the outer surface of said cylindrical wall; said central chamber having present therein a medium which is more compressible than said liquid medium; and said cylindrical wall being made of a predetermined material and having a predetermined thickness such that the cylindrical wall forms a containic membrane having a hoop stiffness which transforms an acoustic pressure wave signal incident to the mandrel and helically wound optical fiber bonded thereto into longitudinal strain variations within said span.
 24. The reflectometer of claim 1, wherein each of: (i) said coherent carrier lightwave signal; (ii) said coherent local oscillator lightwave signal; (iii) said spread spectrum modulated interrogation lightwave signal; (iv) said composite back-propagating lightwave signal; (v) said radio frequency (r.f.) composite difference beat signal; and (vi) each counterpart of said r.f. counterpart of the subcomponents of said second signal component of said composite back-propagating lightwave signal, is a continuous wave (CW) signal.
 25. The reflectometer of claim 1, wherein said continuously reiterated sequences of an autocorrelatable spectrum-spreading signal are binary pseudonoise code sequences.
 26. The reflectometer of claim 25, wherein the pseudonoise code sequences are pseudorandom number (PRN) code sequences.
 27. The reflectometer of claim 1, wherein said continuously reiterated sequences of said autocorrelatable spectrum spreading signals are binary pseudonoise code sequences wherein shifts between binary states of the signal alternatingly shift the radian phase of the carrier substantially between 0° and 180°.
 28. The reflectometer of claim 17, wherein said reference phase signal produced by said fixed frequency oscillator is used in establishing the phase locked relationship between the local oscillator lightwave signal and the carrier lightwave signal.
 29. The reflectometer of claim 10, and: at least one subtracter circuit having first and second inputs for receiving outputs from a selected first and a selected second of said set of n phase demodulators, respectively, to produce a differential phase signal which is representative of the difference between the radian phases of the r.f. counterpart of the said subcomponent of the composite back-propagating lightwave across a region of said fiber optic span having bounds defined by sensing positions corresponding to said selected first and second phase demodulators.
 30. The reflectometer of claim 29, wherein: said at least one subtracter circuit comprises a plurality of subtracter circuits; and a programmable routing and phase signals switching network which routes outputs of selective pairs of said set of n phase demodulators to the inputs of selected subtracter circuits of said plurality to provide a corresponding plurality of outputs at the subtracter circuits of differential phase signals for desired bounds and desired positions of the bounds along said fiber optic span.
 31. A system wherein, at respective sensing stations of a plurality of sensing stations along a span of optical fiber, the system senses input signals of a type having a property of inducing light path changes at regions of the span influenced by such input signals, comprising: means for illuminating an optical fiber span with a CW optical signal; means for retrieving back-propagating portions of the illumination back propagating from a continuum of locations along the span; means for modulating said CW optical signal with a reiterative binary psuedonoise code sequence in a manner which further temporally structures the modulated CW optical signal into a spread spectrum reiterative binary psuedonoise code sequence signal; means for picking off a radio frequency (r.f.) counterpart of the retrieved signal, wherein the r.f. counterpart is in phase locked synchronism with the CW optical signal; means for performing a corresponding plurality of coherent autocorrelation detection processes upon said r.f. counterpart of the retrieved optical signal to conjunctively perform correlation detection and dispreading of the r.f. counterparts in the respective autocorrelation detections of the plurality of autocorrelation detection processes in a corresponding plurality of different timed relationships with respect to the reiterative autocorrelatable form of modulation of the CW optical signal.
 32. Signal sensing apparatus for sensing input signals at an array of a plurality of sensing stations along an optical fiber span, wherein at respective sensing stations of the array the apparatus senses input signals of a type having the property of inducing light path changes within regions influenced by such input signals, said apparatus comprising: an optical wave network comprising a transmitter laser and a lightwave directional coupler, said network being operative to illuminate an optical fiber span with a CW optical signal and to retrieve portions of the illumination back-propagating from a continuum of locations along the fiber span; a modulator operative to modulate and temporally structure the CW optical signal into a CW optical signal with a reiterative spread spectrum form of binary psuedonoise code sequence form of modulated signal; a heterodyner which, in phase locked synchronism with said transmitter laser, receives said retrieved back-propagated portions of illumination and derives therefrom a radio frequency (r.f.) counterpart; and a corresponding plurality of autocorrelation detectors operative to respectively perform coherent correlation processes upon said r.f. counterpart of the retrieved optical signal to conjunctively perform correlation detection and dispreading functions therewith, in respective timed relationships of a corresponding plurality of different timed relationships with respect to said reiterative autocorrelatable form of modulation code. 